Receiving system and method for processing digital broadcast signal in the receiving system

ABSTRACT

A transmitting system, a receiving system, and a method of processing broadcast signals are disclosed. The method for processing a broadcast signal in a broadcast receiver comprises receiving a DTV signal including a data group, the data group including mobile service data, segmented known data sequences, long known data sequences and transmission parameter data, compensating carrier frequency offset of the DTV signal and channel-equalizing the carrier frequency offset compensated DTV signal using at least one of the long known data sequences and segmented known data sequences in the data group of the DTV signal, wherein the channel-equalizing includes performing a Error Correction (FEC) decoding on data located between the segmented known data sequences, and estimating Channel Impulse Response (CIR) using the FEC decoded data as known data.

This application claims the benefit of U.S. Provisional Application No.61/394,789, filed on Oct. 20, 2010 which are hereby incorporated byreference as if fully set forth herein.

BACKGROUND OF THE INVENTION

1. The Field of the Invention

The present invention relates to a digital broadcasting system fortransmitting and receiving a digital broadcast signal, and moreparticularly, to a transmitting system for processing and transmittingthe digital broadcast signal, and a method of processing data in thetransmitting system and the receiving system.

2. Description of the Related Art

The Vestigial Sideband (VSB) transmission mode, which is adopted as thestandard for digital broadcasting in North America and the Republic ofKorea, is a system using a single carrier method. Therefore, thereceiving performance of the digital broadcast receiving system may bedeteriorated in a poor channel environment. Particularly, sinceresistance to changes in channels and noise is more highly required whenusing portable and/or mobile broadcast receivers, the receivingperformance may be even more deteriorated when transmitting mobileservice data by the VSB transmission mode.

SUMMARY OF THE INVENTION

Accordingly, the present invention is directed to a transmitting systemand a method of processing a digital broadcast signal in a transmittingsystem that substantially obviate one or more problems due tolimitations and disadvantages of the related art. An object of thepresent invention is to provide a transmission system which is able totransmit additional mobile service data while simultaneously maintainingthe compatibility with a conventional system for transmitting a digitalbroadcast signal, and a method for processing a broadcast signal.

Another object of the present invention is to provide a transmissionsystem which additionally inserts mobile service data and known datarecognized by an agreement between a transmission system and a receptionsystem into a conventional mobile service data area, thereby enhancingthe reception performance of the mobile service data at the receptionsystem, and a method for processing a broadcast signal.

Another object of the present invention is to provide a transmissionsystem which forms continuous known data sequences by interconnectingdiscontinuous known data belonging to each data group through aconcatenated structure of adjacent data groups, thereby enhancing thereception performance of a broadcast signal at a reception system, and amethod for processing a broadcast signal.

Another object of the present invention is to provide a transmissionsystem which generates information of additional mobile service data byextending signaling information and transmits the generated informationto a reception system, such that the transmission system and thereception end can smoothly communicate with each other, and a method forprocessing a broadcast signal.

A further object of the present invention is to provide a transmittingsystem, a receiving system, and a method for processing broadcastsignals that can enhance the receiving performance of the receivingsystem by performing carrier recovery and channel equalization using theknown data

Additional advantages, objects, and features of the invention will beset forth in part in the description which follows and in part willbecome apparent to those having ordinary skill in the art uponexamination of the following or may be learned from practice of theinvention. The objectives and other advantages of the invention may berealized and attained by the structure particularly pointed out in thewritten description and claims hereof as well as the appended drawings.

To achieve these objects and other advantages and in accordance with thepurpose of the invention, as embodied and broadly described herein, amethod for processing a broadcast signal in a broadcast receivercomprises receiving a DTV signal including a data group, the data groupincluding mobile service data, segmented known data sequences, longknown data sequences and transmission parameter data, compensatingcarrier frequency offset of the DTV signal and channel-equalizing thecarrier frequency offset compensated DTV signal using at least one ofthe long known data sequences and segmented known data sequences in thedata group of the DTV signal, wherein the channel-equalizing includesperforming a Error Correction (FEC) decoding on data located between thesegmented known data sequences, and estimating Channel Impulse Response(CIR) using the FEC decoded data as known data.

The channel-equalizing includes estimating a first CIR using the longknown data sequence, performing FEC decoding the data located betweenthe segmented known data sequences at a decision device, and estimatinga second CIR using the segmented known data sequence and the FEC decodeddata.

The channel-equalizing further includes iterating estimating CIR byusing feedback of the FEC decoded data from the decision device and DTVsignal which is buffered for a period during a processing in thedecision device.

The channel-equalizing further includes performing a pre-equalizing forthe carrier frequency offset compensated DTV signal, performing a FECdecoding on the data located between the segmented known data sequencesof the pre-equalized DTV signal at a decision device, performing abuffering the carrier frequency offset compensated DTV signal during aprocessing of the pre-equalizing and decision device and estimating CIRfor the buffered DTV signal using the FEC decoded data as known data.

The channel-equalizing further includes selecting first CIR of whichvalue is over a threshold value and estimating second CIR using a filterhaving length which is shorter than a channel length and covers thefirst CIR.

Also, an apparatus for processing a digital broadcasting signal in atransmitter is described herein, the apparatus comprises means forfulfilling above mentioned method.

It is to be understood that both the foregoing general description andthe following detailed description of the present invention areexemplary and explanatory and are intended to provide furtherexplanation of the invention as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are included to provide a furtherunderstanding of the invention and are incorporated in and constitute apart of this application, illustrate embodiment(s) of the invention andtogether with the description serve to explain the principle of theinvention. In the drawings:

FIG. 1 is a block diagram illustrating a transmission system accordingto an embodiment of the present invention.

FIG. 2 illustrates a data frame structure for transmitting/receivingmobile data according to an embodiment of the present invention.

FIG. 3 illustrates a structure provided before a data group isinterleaved, when the data group includes 118 mobile service datapackets, according to an embodiment of the present invention.

FIG. 4 illustrates a structure provided after a data group isinterleaved, when the data group includes 118 mobile service datapackets, according to an embodiment of the present invention.

FIG. 5 illustrates regions of a data group including 118 mobile servicedata packets according to an embodiment of the present invention.

FIG. 6 illustrates a data group including (118+M) mobile service datapackets according to an embodiment of the present invention.

FIG. 7 illustrates a table that includes classification of regionscontained in a data group depending on a parade type, transmissionregions of respective ensembles, compatibility, the number of mobileservice data packets contained in one slot, etc. according to anembodiment of the present invention.

FIG. 8 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+M) mobile service datapackets, according to an embodiment of the present invention.

FIG. 9 illustrates a structure provided after a data group isinterleaved, when the data group includes (118+M) mobile service datapackets, according to an embodiment of the present invention.

FIG. 10 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+M) mobile service datapackets, according to an embodiment of the present invention.

FIG. 11 illustrates a structure provided after a data group isinterleaved, when the data group includes (118+M) mobile service datapackets, according to an embodiment of the present invention.

FIG. 12 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

FIG. 13 illustrates a structure provided after a data group isinterleaved when the data group includes (118+38) mobile service datapackets according to an embodiment of the present invention.

FIG. 14 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

FIG. 15 illustrates a structure provided after a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

FIG. 16 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

FIG. 17 illustrates a structure provided after a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

FIG. 18 illustrates a syntax structure of a data field for signalingdigital broadcast data according to an embodiment of the presentinvention.

FIG. 19 is a flowchart illustrating a method for processing andtransmitting digital broadcast data according to an embodiment of thepresent invention.

FIG. 20 is a flowchart illustrating a method for processing andtransmitting digital broadcast data in such a manner that known data isinserted into regions C and D of a data group, when additional mobileservice data is contained in the data group, according to an embodimentof the present invention.

FIG. 21 is a flowchart illustrating a method for processing andtransmitting digital broadcast data in such a manner that mobile servicedata is inserted into a data area for a Reed Solomon (RS) parity and aMoving Picture Experts Group (MPEG) header that are contained in a datagroup, when the data group includes 156 mobile service data packets,according to an embodiment of the present invention.

FIG. 22 is a block diagram illustrating a reception system according toan embodiment of the present invention.

FIG. 23 is a detailed block diagram illustrating a demodulator containedin a channel synchronizer 1301 according to an embodiment of the presentinvention.

FIG. 24 illustrates a known data symbol sequence and a partialcorrelation unit according to an embodiment of the present invention.

FIG. 25 is a conceptual diagram illustrating a method for roughlyestimating an initial frequency offset by dividing a second known datasequence into 8 parts and calculating partial correlation of the 8 partsaccording to an embodiment of the present invention.

FIG. 26 is a conceptual diagram illustrating a method for preciselyestimating a frequency offset using a maximum-likelihood algorithmaccording to an embodiment of the present invention.

FIG. 27 illustrates an example of linear interpolation.

FIG. 28 illustrates an example of linear extrapolation.

FIG. 29 illustrates an example of a channel equalizer according to anembodiment of the present invention.

FIG. 30 illustrates a serial concatenated convolution code (SCCC) codingprocess according to an embodiment of the present invention.

FIG. 31 illustrates a detailed block view showing a block decoderaccording to an embodiment of the present invention.

FIG. 32 is a block diagram illustrating a pattern generator of a symbolinterleaver according to an embodiment of the present invention.

FIG. 33 illustrates an example of a symbol interleaving pattern when anoffset value is set to ‘0’ according to an embodiment of the presentinvention.

FIG. 34 is a conceptual diagram illustrating a process for performingthe symbol interleaving using only a symbol interleaving pattern P(i)according to an embodiment of the present invention.

FIG. 35 illustrates a structure of a Reed Solomon (RS) frame decoderaccording to an embodiment of the present invention.

FIG. 36 illustrates, when an RS frame mode value is equal to ‘00’, anexemplary process of grouping several portions being transmitted to aparade, thereby forming an RS frame and an RS frame reliability map.

FIG. 37 illustrates an example of an error correction decoding processaccording to an embodiment of the present invention.

FIG. 38 illustrates an example of an error correction decoding processaccording to an embodiment of the present invention.

FIG. 39 illustrates a block view of the signaling decoder according toan embodiment of the present invention.

FIG. 40 is a detailed block diagram illustrating an iterative turbodecoder according to an embodiment of the present invention.

FIG. 41( a) illustrates an exemplary case in which a trellis encoder isserially concatenated with an even component encoder, and FIG. 41( b)illustrates an exemplary case in which a trellis encoder is seriallyconcatenated with an odd component encoder.

FIG. 42 is a trellis diagram including states capable of being acquiredwhen a start state for an even decoder is set to ‘00000’.

FIG. 43 is a trellis diagram including states capable of being acquiredwhen a start state for an odd decoder is set to ‘00000’.

FIG. 44 illustrates a detailed embodiment of a process of extracting aTNoG according to an embodiment of the present invention.

FIG. 45 is a diagram showing the form of a known data sequence accordingto an embodiment of the present invention.

FIG. 46 is a diagram showing the structure of a channel equalizeraccording to an embodiment of the present invention.

FIG. 47 is a diagram showing the structure of a channel equalizeraccording to another embodiment of the present invention.

FIG. 48 is a diagram showing an iterative channel equalizer according toan embodiment of the present invention.

FIG. 49 is a diagram showing a channel equalizer including apre-equalizer and a post equalizer according to an embodiment of thepresent invention.

FIG. 50 shows the structure of a decision feedback equalizer of apre-equalizer according to an embodiment of the present invention.

FIG. 51 is a diagram showing the structure of a channel equalizeraccording to another embodiment of the present invention.

FIG. 52 is a diagram showing a CIR estimator according to an embodimentof the present invention.

FIG. 53 shows a CIR estimation method of a known data region accordingto an embodiment of the present invention.

FIG. 54 is a diagram showing a CIR estimation method of a known dataregion according to another embodiment of the present invention.

FIG. 55 is a diagram showing a CIR estimation method of a known dataregion according to another embodiment of the present invention.

FIG. 56 is a diagram showing a method of selecting a sparse window forCIR measurement according to an embodiment of the present invention.

FIG. 57 is a diagram showing a sparse LMS CIR estimator according to anembodiment of the present invention.

FIG. 58 is a diagram showing generation of interpolation error due to asparse window according to an embodiment of the present invention.

FIG. 59 is a diagram showing a method of using information aboutadjacent sparse windows for minimizing error in an interpolation periodaccording to an embodiment of the present invention.

FIG. 60 is a flowchart illustrating a method of receiving and processinga broadcast signal at a receiver according to an embodiment of thepresent invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Reference will now be made in detail to the preferred embodiments of thepresent invention, examples of which are illustrated in the accompanyingdrawings. Wherever possible, the same reference numbers will be usedthroughout the drawings to refer to the same or like parts.

In addition, although the terms used in the present invention areselected from generally known and used terms, some of the termsmentioned in the description of the present invention have been selectedby the applicant at his or her discretion, the detailed meanings ofwhich are described in relevant parts of the description herein.Furthermore, it is required that the present invention is understood,not simply by the actual terms used but by the meaning of each termlying within.

For convenience of description and better understanding of the presentinvention, abbreviations and terms to be use in the present inventionare defined as follows.

Among the terms used in the description of the present invention, mainservice data correspond to data that can be received by a fixedreceiving system and may include audio/video (A/V) data. Morespecifically, the main service data may include A/V data of highdefinition (HD) or standard definition (SD) levels and may also includediverse data types required for data broadcasting. Also, the known datacorrespond to data pre-known in accordance with a pre-arranged agreementbetween the receiving system and the transmitting system.

Additionally, among the terms used in the present invention, “M/H (orMH)” corresponds to the initials of “mobile” and “handheld” andrepresents the opposite concept of a fixed-type system. Furthermore, theM/H service data may include at least one of mobile service data andhandheld service data, and will also be referred to as “mobile servicedata” for simplicity. Herein, the mobile service data not onlycorrespond to M/H service data but may also include any type of servicedata with mobile or portable characteristics. Therefore, the mobileservice data according to the present invention are not limited only tothe M/H service data.

The above-described mobile service data may correspond to data havinginformation, such as program execution files, stock information, and soon, and may also correspond to A/V data. Most particularly, the mobileservice data may correspond to A/V data having lower resolution andlower data rate as compared to the main service data. For example, if anA/V codec that is used for a conventional main service corresponds to aMPEG-2 codec, a MPEG-4 advanced video coding (AVC) or scalable videocoding (SVC) having better image compression efficiency may be used asthe A/V codec for the mobile service. Furthermore, any type of data maybe transmitted as the mobile service data. For example, transportprotocol expert group (TPEG) data for broadcasting real-timetransportation information may be transmitted as the main service data.

Also, a data service using the mobile service data may include weatherforecast services, traffic information services, stock informationservices, viewer participation quiz programs, real-time polls andsurveys, interactive education broadcast programs, gaming services,services providing information on synopsis, character, background music,and filming sites of soap operas or series, services providinginformation on past match scores and player profiles and achievements,and services providing information on product information and programsclassified by service, medium, time, and theme enabling purchase ordersto be processed. Herein, the present invention is not limited only tothe services mentioned above.

Known data—Known data is pre-recognized by an agreement between atransmission system and a reception system, and may be used for channelequalization, etc.

FEC—FEC is an abbreviation of a Forward Error Correction, and is ageneric name of technologies wherein a reception end can spontaneouslycorrect an error of a digital signal transmitted from the transmissionend to the reception end without retransmission of a correspondingsignal by the transmission end.

TPC—TPC is an abbreviation of a Transmission Parameter Channel. TPC iscontained in each data group, and then transmitted. The TPC providesinformation about a data frame and a data group to the reception end,and performs signaling of the provided information.

TS—TS is an abbreviation of a Transport Stream.

RS—RS is an abbreviation of Reed-Solomon.

CRC—CRC is an abbreviation of a Cyclic Redundancy Check.

SCCC—SCCC is an abbreviation of a Serial Concatenated ConvolutionalCode.

PCCC—PCCC is an abbreviation of a Parallel Concatenated ConvolutionalCode.

FIC—FIC is an abbreviation of a Fast information channel. FIC carriescross-layer information. This information primarily includes channelbinding information between ensembles and services.

Embodiments of the present invention will hereinafter be described withreference to the annexed drawings.

FIG. 1 is a block diagram illustrating a transmission system accordingto an embodiment of the present invention.

Referring to FIG. 1, the transmission system includes a packetadjustment unit 101, a pre-processor 102, a data frame encoder 103, ablock processor 104, a signaling encoder 105, a group formatter 106, apacket formatter 107, a Packet multiplexer (Packet MUX) 108, apost-processor 109, a modified data randomizer 110, asystematic/non-systematic RS encoder 111, a data interleaver 112, anon-systematic RS encoder 113, a parity replacer 114, a modified trellisencoder 115, a synchronization multiplexer (Sync MUX) 116, a pilotinserter 117, a VSB modulator 118, and a Radio Frequency (RF)up-converter 119. In addition, the transmission system of FIG. 1 mayfurther include a pre-equalizer filter 120.

When a mobile service data packet and a main service data packet aremultiplexed, there may occur a displacement between a service streampacket including a mobile service stream and another service streampacket including no mobile service stream. In order to compensate forthe displacement, the packet adjustment unit 101 may be used.

The pre-processor 102 configures mobile service data in a form of amobile service structure for transmitting the mobile service data. Inaddition, the pre-processor 102 performs additional FEC coding of mobileservice data. Also, the pre-processor 102 inserts known data. That is,the pre-processor 102 increases the stability of transmission andreception of mobile service data under a mobile environment.

The pre-processor 102 may include a data frame encoder 103, a blockprocessor 103, a block processor 104, a signaling encoder 105, a groupformatter 106, a packet formatter 107, and a packet multiplexer (packetMUX) 108. In other words, the above-mentioned constituent components maybe contained in the pre-processor 102, and may be configured separatelyfrom the pre-processor 102.

The data frame encoder 103 randomizes mobile service data, and performsRS encoding and CRC encoding of the mobile service data.

The block processor 104 converts an RS frame portion into an SCCC block.The block processor 104 converts a mobile service data byte contained inthe SCCC block into bit-based mobile service data. The block processor104 performs convolution encoding of ½, ⅓, or ¼ rate on the bit-basedmobile service data. In this case, the ½ rate means an encoding processin which two bits are output in response to an input of one bit, the ⅓rate means an encoding process in which three bits are output inresponse to an input of two bits, and the ¼ rate means an encodingprocess in which four bits are output in response to an input of fourbits. Output bits are contained in a symbol. The block processor 104performs interleaving of the convolution-encoded output symbol. Theblock processor 104 converts an interleaved symbol into byte-based data,and converts an SCCC block into a data block. A detailed description ofthe data block will hereinafter be described in detail.

The signaling encoder 105 generates signaling information for signalingat a reception end, performs FEC encoding and PCCC encoding of thegenerated signaling information, and inserts the signaling informationinto some regions of the data group. For example, examples of thesignaling information may be a transmission parameter channel (TPC)data, fast information channel (FIC) data, and the like.

The group formatter 106 forms a data group using the output data of theblock processor 104. The group formatter 106 maps FEC-encoded mobileservice data to an interleaved form of a data group format. At thistime, the above-mentioned mapping is characterized in that FEC-encodedmobile service data is inserted into either a data block of acorresponding group or a group region according to a coding rate of eachFEC-encoded mobile service data received from the block processor 104.In addition, the group formatter 106 inserts signaling data, a data byteused for initializing the trellis encoder, and a known data sequence.Further, the group formatter 106 inserts main service data, and aplace-holder for an MPEG-2 header and a non-systematic RS parity. Thegroup formatter 106 may insert dummy data to generate a data group of adesired format. After inserting various data, the group formatter 106performs deinterleaving of data of the interleaved data group. Afterperforming the deinterleaving operation, the data group returns to anoriginal group formed before the interleaving operation.

The packet formatter 107 converts output data of the group formatter 106into a Transport Stream (TS) packet. In this case, the TS packet is amobile service data packet. In addition, the output of the packetformatter 107 according to an embodiment of the present invention ischaracterized in that it includes (118+M) mobile service data packets ina single data group. In this case, M is 38 or less.

The packet multiplexer (Packet MUX) 108 multiplexes a packet includingmobile service data processed by the pre-processor 102 and a packetincluding main service data output from the packet adjustment unit 101.In this case, the multiplexed packet may include (118+M) mobile servicedata packets and L main service data packets. For example, according toan embodiment of the present invention, M is any one of integers from 0to 38, and the sum of M and L is set to 38. In other words, although thepacket multiplexer (packet MUX) 108 may multiplex the mobile servicedata packet and the main service data packet, in the case where thenumber of input main service data packets is set to ‘0’ (i.e., L=0),only the mobile service data packet is processed by the packetmultiplexer (packet MUX) 108, such that the packet multiplexer (packetMUX) 108 outputs the processed mobile service data packet only.

The post-processor 109 processes mobile service data in such a mannerthat the mobile service data generated by the present invention can bebackward compatible with a conventional broadcast system. In accordancewith one embodiment of the present invention, the post-processor 109 mayinclude a modified data randomizer 110, a systematic/non-systematic RSencoder 111, a data interleaver 112, a non-systematic RS encoder 113, aparity replacer 114 and a modified trellis encoder 115. In other words,each of the above-mentioned constituent components may be locatedoutside of the post-processor 109 according to the intention of adesigner as necessary.

The modified data randomizer 110 does not perform randomizing of amobile service TS packet, and bypasses a mobile service TS packet. Themodified data randomizer 110 performs randomizing of the main servicedata TS packet. Therefore, according to one embodiment of the presentinvention, the randomizing operation is not performed when a data groupgenerated by the pre-processor 102 has no main service data.

In the case where input data is a main service data packet, thesystematic/non-systematic RS encoder 111 performs systematic RS encodingof the main service data packet acting as the input data, such that itgenerates RS FEC data. In the case where input data is a mobile servicedata packet, the systematic/non-systematic RS encoder 111 performsnon-systematic RS encoding, such that it generates RS FEC data. Inaccordance with one embodiment of the present invention, thesystematic/non-systematic RS encoder 111 generates RS FEC data havingthe size of 20 bytes during the systematic/non-systematic RS encodingprocess. The RS FEC data generated in the systematic RS encoding processis added to the end of a packet having the size of 187 bytes. RS FECdata generated in the non-systematic RS encoding process is insertedinto the position of an RS parity byte predetermined in each mobileservice data packet. Therefore, according to one embodiment of thepresent invention, in the case where the data group generated by thepre-processor has no main service data, the systematic RS encoder 111for main service data performs no RS encoding. In this case, thenon-systematic RS encoder 111 does not generate a non-systematic RSparity for backward compatibility.

The data interleaver 112 performs byte-based interleaving of data thatincludes main service data and mobile service data.

In the case where it is necessary to initialize the modified trellisencoder 115, the non-systematic RS encoder 113 receives an internalmemory value of the modified trellis encoder 115 as an input, andreceives mobile service data from the data interleaver 112 as an input,such that it changes initialization data of mobile service data to amemory value. The non-systematic RS encoder 113 performs non-systematicRS encoding of the changed mobile service data, and outputs thegenerated RS parity to the parity replacer 114.

In the case where it is necessary to initialize the modified trellisencoder 115, the parity replacer 114 receives mobile service data outputfrom the data interlever 112, and replaces an RS parity of the mobileservice data with an RS parity generated from the non-systematic RSencoder 113.

In the case where the data group generated in the pre-processor does notinclude main service data at all, the data group need not have an RSparity for backward compatibility. Accordingly, in accordance with oneembodiment of the present invention, the non-systematic RS encoder 113and the parity replacer 114 do not perform each of the above-mentionedoperations, and bypass corresponding data.

The modified trellis encoder 115 performs trellis encoding of outputdata of the data interleaver 112. In this case, in order to allow dataformed after the trellis encoding to have known data pre-engaged betweena transmission end and a reception end, a memory contained in themodified trellis encoder 115 should be initialized before the beginningof the trellis encoding. The above-mentioned initialization operationbegins by trellis initialization data belonging to a data group.

The synchronization multiplexer (Sync MUX) 116 inserts a fieldsynchronization signal and a segment synchronization signal into outputdata of the modified trellis encoder 115, and multiplexes the resultantdata.

The pilot inserter 117 receives the multiplexed data from thesynchronization multiplexer (Sync MUX) 116, and inserts a pilot signal,that is used as a carrier phase synchronization signal for demodulatinga channel signal at a reception end, into the multiplexed data.

The VSB modulator 118 performs VSB modulation so as to transmit data.

The transmission unit 119 performs frequency up-conversion of themodulated signal, and transmits the resultant signal.

In the present invention, the transmitting system provides backwardcompatibility in the main service data so as to be received by theconventional receiving system. Herein, the main service data and themobile service data are multiplexed to the same physical channel andthen transmitted.

Furthermore, the transmitting system according to the present inventionperforms additional encoding on the mobile service data and inserts thedata already known by the receiving system and transmitting system(e.g., known data), thereby transmitting the processed data.

Therefore, when using the transmitting system according to the presentinvention, the receiving system may receive the mobile service dataduring a mobile state and may also receive the mobile service data withstability despite various distortion and noise occurring within thechannel.

FIG. 2 illustrates a data frame structure for transmitting/receivingmobile service data according to one embodiment of the presentinvention.

In the embodiment of the present invention, the mobile service data arefirst multiplexed with main service data in DATA frame units and, then,modulated in a VSB mode and transmitted to the receiving system.

The term “data frame” mentioned in the embodiment of the presentinvention may be defined as the concept of a time during which mainservice data and mobile service data are transmitted. For example, onedata frame may be defined as a time consumed for transmitting 20 VSBdata frames.

At this point, one DATA frame consists of K1 number of sub-frames,wherein one sub-frame includes K2 number of slots. Also, each slot maybe configured of K3 number of data packets. In the embodiment of thepresent invention, K1 will be set to 5, K2 will be set to 16, and K3will be set to 156 (i.e., K1=5, K2=16, and K3=156). The values for K1,K2, and K3 presented in this embodiment either correspond to valuesaccording to a preferred embodiment or are merely exemplary. Therefore,the above-mentioned values will not limit the scope of the presentinvention.

In the example shown in FIG. 2, one DATA frame consists of 5 sub-frames,wherein each sub-frame includes 16 slots. In this case, the DATA frameaccording to the present invention includes 5 sub-frames and 80 slots.

Also, in a packet level, one slot is configured of 156 data packets(i.e., transport stream packets), and in a symbol level, one slot isconfigured of 156 data segments. Herein, the size of one slotcorresponds to one half (½) of a VSB field. More specifically, since one207-byte data packet has the same amount of payload data as payload dataof a segment, a data packet prior to being interleaved may also be usedas a data segment.

156 data packets contained in a slot may be composed of 156 main servicedata packets, may be composed of 118 mobile service data packets and 38main service data packets, or may be composed of (118+M) mobile servicedata packets and L main service data packets. In this case, the sum of Mand L may be set to 38 according to one embodiment of the presentinvention. In addition, M may be zero ‘0’ or a natural number of 38 orless.

One data group is transmitted during a single slot. In this case, thetransmitted data group may include 118 mobile service data packets or(118+M) mobile service data packets.

That is, a data group may be defined as a set of data units includingmobile service data present in one slot. In this case, the mobileservice data may be defined as pure mobile service data, or may bedefined as the concept that includes data for transmitting mobileservice data, such as signaling data, known data, etc.

FIG. 3 illustrates a structure acquired before a data group isinterleaved, when the data group includes 118 mobile service datapackets, according to an embodiment of the present invention.

Referring to FIG. 3, the data group includes 118 TS packets that includeat least one of FEC-encoded mobile service data, MPEG header, trellisinitialization data, known data, signaling data, RS parity data anddummy data. For convenience of description and better understanding ofthe present invention, a TS packet contained in the data group isdefined as a mobile service data packet according to the presentinvention.

The data group shown in FIG. 3 includes 118 mobile service data packets,such that it can be recognized that the slot via which theabove-mentioned data group is transmitted is used for transmitting 38main service data packets.

FIG. 4 illustrates a structure acquired after a data group isinterleaved, when the data group includes 118 mobile service datapackets, according to an embodiment of the present invention.

Referring to FIG. 4, the data group including 118 mobile service datapackets is interleaved such that a data group including 170 segments isformed.

In this case, the above-mentioned example in which 118 mobile servicedata packets are distributed to 170 segments has been disclosed only forillustrative purposes and better understanding of the present invention.The number of data segments formed after the data group is interleavedmay be changed to another according to the degree of interleaving.

FIG. 4 shows an example of dividing a data group prior to beingdata-interleaved into 10 DATA blocks (i.e., DATA block 1 (B1) to DATAblock 10 (B10)). In other word, DATA Block can be defined as atransmission block containing mobile service data or main and mobileservice data in segment level. In this example, each DATA block has thelength of 16 segments. Referring to FIG. 4, only the RS parity data areallocated to a portion of 5 segments before the DATA block 1 (B1) and 5segments behind the DATA block 10 (B10). The RS parity data are excludedin regions A to D of the data group.

More specifically, when it is assumed that one data group is dividedinto regions A, B, C, and D, each DATA block may be included in any oneof region A to region D depending upon the characteristic of each DATAblock within the data group. At this point, according to an embodimentof the present invention, each DATA block may be included in any one ofregion A to region D based upon an interference level of main servicedata.

Herein, the data group is divided into a plurality of regions to be usedfor different purposes. More specifically, a region of the main servicedata having no interference or a very low interference level may beconsidered to have a more resistant (or stronger) receiving performanceas compared to regions having higher interference levels. Additionally,when using a system inserting and transmitting known data in the datagroup, wherein the known data are known based upon an agreement betweenthe transmitting system and the receiving system, and when consecutivelylong known data are to be periodically inserted in the mobile servicedata, the known data having a predetermined length may be periodicallyinserted in the region having no interference from the main service data(i.e., a region wherein the main service data are not mixed). However,due to interference from the main service data, it is difficult toperiodically insert known data and also to insert consecutively longknown data to a region having interference from the main service data.

Referring to FIG. 4, DATA block 4 (B4) to DATA block 7 (B7) correspondto regions without interference of the main service data. DATA block 4(B4) to DATA block 7 (B7) within the data group shown in FIG. 4correspond to a region where no interference from the main service dataoccurs. In this example, a long known data sequence is inserted at boththe beginning and end of each DATA block. In the description of thepresent invention, the region including DATA block 4 (B4) to DATA block7 (B7) will be referred to as “region A (=B4+B5+B6+B7)”. As describedabove, when the data group includes region A having a long known datasequence inserted at both the beginning and end of each DATA block, thereceiving system is capable of performing equalization by using thechannel information that can be obtained from the known data. Therefore,the strongest equalizing performance may be yielded (or obtained) fromone of region A to region D.

In the example of the data group shown in FIG. 4, DATA block 3 (B3) andDATA block 8 (B8) correspond to a region having little interference fromthe main service data. Herein, a long known data sequence is inserted inonly one side of each DATA block B3 and B8. More specifically, due tothe interference from the main service data, a long known data sequenceis inserted at the end of DATA block 3 (B3), and another long known datasequence is inserted at the beginning of DATA block 8 (B8). In thepresent invention, the region including DATA block 3 (B3) and DATA block8 (B8) will be referred to as “region B (=B3+B8)”. As described above,when the data group includes region B having a long known data sequenceinserted at only one side (beginning or end) of each DATA block, thereceiving system is capable of performing equalization by using thechannel information that can be obtained from the known data. Therefore,a stronger equalizing performance as compared to region C/D may beyielded (or obtained).

Referring to FIG. 4, DATA block 2 (B2) and DATA block 9 (B9) correspondto a region having more interference from the main service data ascompared to region B. A long known data sequence cannot be inserted inany side of DATA block 2 (B2) and DATA block 9 (B9). Herein, the regionincluding DATA block 2 (B2) and DATA block 9 (B9) will be referred to as“region C (=B2+B9)”.

Finally, in the example shown in FIG. 4, DATA block 1 (B1) and DATAblock 10 (B10) correspond to a region having more interference from themain service data as compared to region C. Similarly, a long known datasequence cannot be inserted in any side of DATA block 1 (B1) and DATAblock 10 (B10).

Referring to FIG. 4, it can be readily recognized that the regions A andB of the data group includes signaling data used for signaling at areception end.

FIG. 5 illustrates regions of a data group including 118 mobile servicedata packets according to an embodiment of the present invention.

FIG. 5 is shown for better understanding of individual regions of thedata group shown in FIG. 4.

As can be seen from FIG. 5, the data group including 118 mobile servicedata packets can be divided into four regions A, B, C and D. The Aregion is located at the center of the data group, and the B region islocated at the exterior of the A region using the A region as areference line. The C region is located at the exterior of the B regionon the basis of the A and B regions. The D region is located at theexterior of the C area on the basis of the A, B, and C regions.

FIG. 6 illustrates a data group including (118+M) mobile service datapackets according to an embodiment of the present invention.

Referring to FIG. 6( a), the data group includes A, B, C, D and Eregions. The data group is contained in a slot including 156 packets.That is, a predetermined number of packets contained in one slot formthe data group, and such packets include mobile service data.

After 118 mobile service data packets fixed in the data group areinterleaved, the data group is divided into A, B, C and D regions asshown in FIG. 4.

Meanwhile, a variable number (M) of mobile service data packets capableof being contained in the data group are contained in an additionalregion E. In the case where the data group in one slot is composed of118 mobile service data packets, the E region can be defined as aspecific region acquired when mobile service data packets are added tothe region composed of only main service data packets. In other words,the E region may include a scalable number of mobile service datapackets in one slot.

The mapping format of the mobile service data packets in the E regionmay be changed according to the intention of a designer. In other words,according to one embodiment of the present invention, when the number ofmobile service data packets is 38 or less (i.e., M<38) as shown in FIG.6( a), a specific packet region in one slot remains empty in such amanner that the empty specific packet region can be used as a mainservice data packet region, and therefore mobile service data packetscan be mapped to the remaining parts. According to another embodiment ofthe present invention, mobile service data packets can be mapped to thedata group in such a manner that M scalable mobile service data packetscontained in the E region are spaced apart from one another at intervalsof a predetermined distance.

FIG. 6( b) illustrates a structure acquired after the data groupincluding the E region as shown in FIG. 6( a) is interleaved.

Referring to FIG. 6( b), 10 blocks (B1˜B10) contained in the data groupform A, B, C and D regions using the same pattern as in the data groupshown in FIG. 5. However, the E region including M scalable mobileservice data packets is formed as an additional block.

As can be seen from FIG. 6( b), the E region belonging to the data groupmay be contained in a plurality of blocks, and respective blocks maycorrespond to a scalable number of VSB segments. Mobile service dataadditionally transmitted through the E region is distributed to 4 or 5blocks.

Meanwhile, in the case where the data group does not include mainservice data, the E region includes a block which includes an area of aplace-holder that includes not only an RS parity but also an MPEG headerfor backward compatibility with a conventional digital broadcast system.In other words, in the case where the data group does not include mainservice data, the RS parity and the MPEG header for backwardcompatibility need not be used, such that an area reserved for the RSparity and the MPEG header is allocated to an area for mobile servicedata and forms a block contained in the E region.

Although 5 blocks are contained in the E region as shown in FIG. 6( b),the scope or spirit of the present invention is not limited onlythereto. That is, the number of segments contained in each block of theE region may be scalable, such that the number of blocks contained inthe E regions may also be scalable.

In the meantime, according to the present invention, the E regioncontained in the data group is determined by M scalable mobile servicedata packets, such that an appropriate number of mobile service datapackets can be transmitted according to an amount of mobile service datato be transmitted, resulting in an increased transmission efficiency.

In addition, additional mobile service data packets are transmittedthrough the E regain of the data group, such that the demand of a userwho desires to use a high-quality mobile service that requires a largeamount of data can be satisfied.

FIG. 7 illustrates a table that includes classification of regionscontained in a data group depending on a parade type, transmissionregions of respective ensembles, compatibility, the number of mobileservice data contained in one slot, etc. according to an embodiment ofthe present invention.

For convenience of description and better understanding of the presentinvention, specific information indicating the number of mobile servicedata packets contained in one slot is defined as a scalable mode. Sincea data group including mobile service data is transmitted during oneslot, the scalable mode may represent information indicating the numberof mobile service data packets belonging to the data group. In otherwords, in the case where the data group contained in one slot includes(118+M) mobile service data packets and L main service data packets(where L is the number of remaining packets), it may be considered thatthe scalable mode includes M or L number of information. In other words,the scalable mode may represent specific information indicating at leastone of M number of information and ‘156−(118+M)’ number of information(hereinafter ‘156−(118+M)’ is referred to as ‘L’).

In accordance with one embodiment of the present invention, M may beequal to or higher than ‘0’, or may be an integer of 38 or less. Forexample, M may be set to 11, 20, 29 or 38.

Meanwhile, in this embodiment, a collection of services is defined byconcept of ensemble. One ensemble has the same QoS, and is coded withthe same FEC code. Also, the ensemble has unique identifier (i.e.,ensemble id), and is a collection of consecutive RS frames having thesame FEC code. In other words, a set of RS frames logically forms anEnsemble.

Meanwhile, according to one embodiment of the present invention, an RSframe is defined as a two-dimensional data frame. Mobile service data istransmitted through the RS frame.

Referring to FIG. 7, in the processing and transmitting scheme fortransmitting a digital broadcast signal according to one embodiment ofthe present invention, a scalable mode for use in the first typerepresents that M is set to ‘0’. In other words, the scalable moderepresents that the number of mobile service data packets contained inthe data group is 118. One ensemble can be transmitted through A, B, Cand D regions. In this case, a single ensemble is defined as a primaryensemble.

In the case of a second type, a scalable mode represents that M is setto ‘0’ in the same manner as in the first type. However, two ensemblesare transmitted through the A, B, C and D regions. At this time, theensemble transmitted through the A and B regions is referred to as aprimary ensemble, and the ensemble transmitted through the C and Dregions is referred to as a secondary ensemble. The primary ensemble andthe secondary ensemble have different quality of services (QoSs) anddifferent FEC codes, such that the embodiment of the present inventionassumes that the primary ensemble and the secondary ensemble transmitdifferent usages of mobile service data.

In the case of a third type, a scalable mode represents that M is notset to ‘0’. That is, the third type represents that not only 118 mobileservice data packets but also additional mobile service data packets arecontained in the data group. Although the embodiment of the presentinvention assumes that M is set to 30, the scope or spirit of thepresent invention is not limited only thereto because M is a scalablenumber of 38 or less. In accordance with the embodiment of the presentinvention, the primary ensemble is transmitted through the A, B, C and Dregions, and the secondary ensemble is transmitted through the E regionadded to the data group in response to a value of M.

In the case of a fourth type, a scalable mode is identical to that ofthe third type. However, the primary ensemble is transmitted through theA and B regions, and the secondary ensemble is transmitted through theC, D and E regions. At this time, while the A and B regions maintain thecompatibility with a current digital mobile broadcast system, the C, Dand E regions support additional payload for mobile service data.

In the case of a fifth type, a scalable mode is identical to those ofthird and fourth types. However, the primary ensemble is transmittedthrough all of the A, B, C, D and E regions.

In the case of a sixth type, a scalable mode represents that M is set to38. That is, the data group including 156 mobile service data packets inone slot is transmitted. The primary ensemble is transmitted through theA, B, C and D regions, and the secondary ensemble is transmitted throughthe E region.

In the case of a seventh type, a scalable mode is identical to that ofthe sixth type. However, the primary ensemble is transmitted through theA and B regions, and the secondary ensembles are transmitted through theC, D and E regions. Therefore, while the A and B regions maintain thecompatibility with a current digital mobile broadcast system, the C, Dand E regions support additional payload for mobile service data.

In the case of an eighth type, a scalable mode is identical to those ofthe sixth and seventh types. However, the primary ensemble istransmitted through all of the A, B, C, D and E regions.

In accordance with one embodiment of the present invention, the regionvia which the primary ensemble and the secondary ensemble aretransmitted is defined as an RS frame mode, can be identified at thereception end. That is, the RS frame mode may be defined as specificinformation that indicates either a first case in which the primaryensemble is transmitted through the A and B regions and the secondaryensemble is transmitted through the C, D and E regions, or a second casein which the primary ensemble is transmitted through the A, B, C, D andE regions.

In accordance with one embodiment of the present invention, in the casewhere the scalable mode represents that M is set to 38, or in the casewhere the number of mobile service data packets contained in the datagroup is 156, no main service data packets are contained in the slot.Therefore, the RS parity and the MPEG header for backward compatibilityneed not be inserted into the data group. In this case, mobile servicedata can be inserted into a reserved data area for both the RS parityand the MPEG header for backward compatibility. At this time, thedigital broadcast transmitter can transmit much more mobile servicedata.

In accordance with still another embodiment of the present invention,the data groups respectively transmitted through 16 slots contained inone sub-frame may have different scalable modes. Hence, even when thescalable mode of the data group wherein a specific slot is contained inone sub-frame represents that M is set to 38, some scalable modes fromamong the data groups of the remaining slots represent that M is set toanother value not ‘38’. In this case, the corresponding sub-frameincludes main service data, such that each data group contained in thecorresponding sub-frame should include the RS parity and the MPEGheader. That is, in this case, it is impossible to insert mobile servicedata into a data area reserved for both the RS parity and the MPEGheader for backward compatibility. However, it is possible to maintainthe compatibility with a conventional digital broadcast system.

FIG. 8 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+M) mobile service datapackets, according to an embodiment of the present invention.

Referring to FIG. 8, the data group includes mobile service data of theA and B regions, mobile service data of the C and D regions, mobileservice data of the E region, an MPEG header, trellis initializationdata, known data, signaling data, RS parity data, and dummy data.

Referring to the data group of FIG. 8, M is denoted by an integer of‘0<M<38’. In other words, the number of mobile service data packetscontained in the data group is higher than 118 and is lower than 156.Although FIG. 8 illustrates that M is set to 30 as an example, M is notlimited only thereto and may also be set to another value as necessary.

Referring to FIG. 8, a TS packet region not allocated to a mobileservice data packet from among 156 TS packets belonging to one slot isreserved for main service data.

FIG. 9 illustrates a structure provided after a data group isinterleaved, when the data group includes (118+M) mobile service datapackets, according to an embodiment of the present invention.

The structure shown in FIG. 9 is identical to a structure formed afterthe data group of FIG. 8 is interleaved.

As can be seen from the data group shown in FIG. 9, the primary ensembleis transmitted through the A, B, C and D regions of the data group, andthe secondary ensemble is transmitted through the E region of the datagroup. The mobile service data of the A, B, C and D regions maintain thecompatibility with a conventional digital mobile broadcast system.

Although the data group of FIG. 9 is divided into 10 blocks belonging tothe A, B, C and D regions and two additional blocks belonging to the Eregion, the number of blocks belonging to the E block is not limitedonly to ‘2’ and may be changed to another number not ‘2’ according tothe intention of a designer.

As shown in FIG. 9, known data is inserted into the E region, such thatthe reception performance of the reception end is increased in the Eregion.

In this case, the term “known data” means specific data pre-engagedbetween the transmission end and the reception end. The transmission endtransmits the pre-engaged data as known data to the reception end.Signal distortion may occur in data transmission. In this case, thereception end may perform channel equalization and the like by referringto the degree of distortion of known data transmitted from thetransmission end. In other words, the known data is adapted to increasethe reception performance of a broadcast signal at the reception end.

FIG. 10 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+M) mobile service datapackets, according to an embodiment of the present invention.

Referring to FIG. 10, the data group includes mobile service data of theA and B regions, mobile service data of the C and D regions, mobileservice data of the E region, an MPEG header, trellis initializationdata, known data, signaling data, RS parity data, and dummy data.

Referring to the data group of FIG. 10, M is denoted by an integer of‘0<M<38’. In other words, the number of mobile service data packetscontained in the data group is higher than 118 and is lower than 156.Although FIG. 10 illustrates that M is set to 30 as an example, M is notlimited only thereto and may also be set to another value as necessary.

FIG. 11 illustrates a structure provided after a data group isinterleaved, when the data group includes (118+K) mobile service datapackets, according to an embodiment of the present invention.

The structure shown in FIG. 11 is identical to a structure formed afterthe data group of FIG. 10 is interleaved.

As can be seen from the data group shown in FIG. 11, the primaryensemble is transmitted through the A and B regions of the data group,and the secondary ensemble is transmitted through the C, D and E regionsof the data group. The mobile service data of the A and B regionsmaintain the compatibility with a conventional digital mobile broadcastsystem.

In addition, the primary ensemble may also be transmitted through all ofthe A, B, C, D and E regions.

Although the data group of FIG. 11 is divided into 10 blocks belongingto the A, B, C and D regions and two additional blocks belonging to theE region, the number of blocks belonging to the E block is not limitedonly to ‘2’ and may be changed to another number not ‘2’ according tothe intention of a designer.

Referring to FIG. 11, known data 1101 is additionally inserted into theC and D regions in addition to the A and B regions. Differently from thedata group shown in FIG. 4 or FIG. 9, additional known data is insertedinto the C and D regions, such that the reception performance of mobileservice data transmitted through the C and D regions can be increased.

FIG. 12 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

Referring to FIG. 12, the data group includes mobile service data of theA and B regions, mobile service data of the C and D regions, mobileservice data of the E region, an MPEG header, trellis initializationdata, known data, signaling data, RS parity data, and dummy data.

As shown in FIG. 12, the E region has no main service data packets, suchthat the region for the RS parity and the MPEG header is not present inthe E region. Therefore, the above-mentioned regions may be adapted totransmit mobile service data, such that much more mobile service datacan be transmitted.

FIG. 13 illustrates a structure provided after a data group isinterleaved when the data group includes (118+38) mobile service datapackets according to an embodiment of the present invention.

The structure shown in FIG. 13 is identical to a structure formed afterthe data group of FIG. 12 is interleaved.

As can be seen from the data group shown in FIG. 13, the primaryensemble is transmitted through the A, B, C and D regions of the datagroup, and the secondary ensemble is transmitted through the E region ofthe data group. Since the A, B, C and D regions are identical to thoseof a conventional data group, they can maintain the compatibility with aconventional digital mobile broadcast system. In addition, additionalmobile service data can be transmitted through the E region.

Although the data group of FIG. 13 is divided into 10 blocks belongingto the A, B, C and D regions and two additional blocks belonging to theE region, the number of blocks belonging to the E block is not limitedonly to ‘2’ and may be changed to another number not ‘2’ according tothe intention of a designer.

Referring to FIG. 13, known data is inserted into the E region.Therefore, the reception performance of the reception end is increasedin the E region. As described above, mobile service data is insertedinto the reserved area for both the RS parity and the MPEG headerpresent in the E region, such that much more mobile service data can betransmitted.

FIG. 14 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

Referring to FIG. 14, the data group includes mobile service data of theA and B regions, mobile service data of the C and D regions, mobileservice data of the E region, an MPEG header, trellis initializationdata, known data, signaling data, RS parity data, and dummy data.

As shown in FIG. 14, the C, D and E regions do not include main servicedata packets and the region for an RS parity and an MPEG header.Therefore, the above-mentioned regions may be adapted to transmit mobileservice data, such that much more mobile service data can betransmitted.

FIG. 15 illustrates a structure provided after a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

The structure shown in FIG. 15 is identical to a structure formed afterthe data group of FIG. 14 is interleaved.

As can be seen from the data group shown in FIG. 15, the primaryensemble is transmitted through the A and B regions of the data group,and the secondary ensemble is transmitted through the C, D and E regionof the data group. Since the A and B regions include the RS parity andthe MPEG header, they can maintain the compatibility with a conventionaldigital mobile broadcast system.

Although the data group of FIG. 15 is divided into 10 blocks belongingto the A, B, C and D regions and two additional blocks belonging to theE region, the number of blocks belonging to the E block is not limitedonly to ‘2’ and may be changed to another number not ‘2’ according tothe intention of a designer.

Referring to FIG. 15, additional known data 15001 is inserted into the Cand D regions in addition to the A and B regions. The data group shownin FIG. 15 is not affected by main service data, such that successiveknown data sequences can be contained in the C and D regions differentlyfrom the data group shown in FIG. 11. Therefore, the receptionperformance of mobile service data transmitted through the C and Dregions at the reception end can be greatly increased.

In accordance with the present invention, the number of known datasequences inserted into the C and D regions is not limited only to aspecific number. Therefore, according to the intention of a designer, aproper number of known data sequences required for enhancing thereception performance of the reception end can be inserted. Inaccordance with one embodiment of the present invention, 3 known datasequences are inserted into the C region, and 2 known data sequences areinserted into the D region.

In addition, as described above, mobile service data is inserted into anarea reserved for the RS parity and the MPEG header in the C, D and Eregions, such that much more mobile service data can be transmitted.

FIG. 16 illustrates a structure provided before a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

As for the data group shown in FIG. 16, on the condition that 16 slotscontained in one sub-frame transmit a data group including 156 mobileservice data packets, the data group of FIG. 16 may represent any one ofdata group types.

The data group shown in FIG. 16 includes mobile service data of the Aand B regions, mobile service data of the C and D regions, mobileservice data of the E region, trellis initialization data, known data,signaling data, and dummy data. That is, the data group of FIG. 16 doesnot include the RS parity and the MPEG header for backwardcompatibility.

As shown in FIG. 16, the A, B, C, D and E regions do not include theregion for the RS parity and the MPEG header. Therefore, theabove-mentioned regions can be used to transmit mobile service data,such that much more mobile service data can be transmitted.

FIG. 17 illustrates a structure provided after a data group isinterleaved, when the data group includes (118+38) mobile service datapackets, according to an embodiment of the present invention.

The structure shown in FIG. 17 is identical to a structure formed afterthe data group of FIG. 16 is interleaved.

Although the data group of FIG. 17 is divided into 10 blocks belongingto the A, B, C and D regions and two additional blocks belonging to theE region, the number of blocks belonging to the E block is not limitedonly to ‘2’ and may be changed to another number not ‘2’ according tothe intention of a designer.

Referring to FIG. 17, additional known data is inserted into the C and Dregions in addition to the A and B regions. The data group shown in FIG.17 is not affected by main service data, such that successive known datasequences can be contained in the C and D regions differently from thedata group shown in FIG. 11. Therefore, the reception performance ofmobile service data transmitted through the C and D regions at thereception end can be greatly increased.

In addition, first known data present in the E region of the first datagroup may be connected to second known data present in the upper C and Dregions of the second data group that is adjacent to the first datagroup. In this case, a known data sequence may be assigned to an overallarea of the data group. As a result, the reception performance of mobileservice data in the case of using the overall area of the group ishigher than the reception performance of mobile service data in anothercase of using a conventional data group.

In accordance with another embodiment of the present invention, whenknown data of the first data group is connected to known data of thesecond group that is adjacent to the first data group, known datainstead of trellis initialization data inserted in the front end of eachknown data may be additionally inserted. In this case, the trellisinitialization data to be located at the front end of the connectedknown data sequence should be contained in the data group.

In addition, as shown in FIG. 17, in the A, B, C, D and E regions,mobile service data is inserted into the reserved area for the RS parityand the MPEG header, such that much more mobile service data can betransmitted within one data group.

FIG. 18 illustrates a syntax structure of a data field for signalingdigital broadcast data according to an embodiment of the presentinvention.

For convenience of description and better understanding of the presentinvention, the above-mentioned data field will hereinafter be referredto as a Transmission Parameter Channel (TPC).

The TPC data may include a sub-frame_number field, a slot_number field,a parade_id field, a starting_group_number (SGN) field, anumber_of_groups (NoG) field, a parade_repetition_cycle (PRC) field, anRS_frame_mode field, an RS_code_mode_primary field, anRS_code_mode_secondary field, an SCCC_block_mode field, anSCCC_outer_code_mode_A field, an SCCC_outer_code_mode_B field, anSCCC_outer_code_mode_C field, an SCCC_outer_code_mode_D field, anFIC_version field, a parade_continuity_counter field, a TNoG field and aTPC_protocol_version field.

The Sub-Frame_number field shall be the current Sub-Frame number withinthe DATA Frame, which is transmitted for DATA Frame synchronization. Itsvalue shall range from 0 to 4

The Slot_number field is the current Slot number within the Sub-Frame,which is transmitted for DATA Frame synchronization. Its value shallrange from 0 to 15

The Parade_id field identifies the Parade to which this Group belongs.The value of this field may be any 7-bit value. Each Parade in a DATAtransmission shall have a unique Parade_id. Communication of theParade_id between the physical layer and the management layer shall beby means of an Ensemble_id formed by adding one bit to the left of theParade_id. If the Ensemble_id is for the primary Ensemble deliveredthrough this Parade, the added MSB shall be ‘0’. Otherwise, if it is forthe secondary Ensemble, the added MSB shall be ‘1’.

The starting_Group_number (SGN) field shall be the first Slot_number fora Parade to which this Group belongs.

The number_of_Groups (NoG) field shall be the number of Groups in aSub-Frame assigned to the Parade to which this Group belongs, minus 1,e.g., NoG=0 implies that one Group is allocated to this Parade in aSub-Frame

The Parade_repetition_cycle (PRC) field shall be the cycle time overwhich the Parade is transmitted, minus 1, specified in units of DATAFrames.

The RS_Frame_mode field represents that one parade transmits one RSframe or two RS frames.

The RS_code_mode_primary field shall be the RS code mode for the primaryRS frame.

The RS_code_mode_secondary field shall be the RS code mode for thesecondary RS frame.

The SCCC_Block_mode field represents how DATA blocks within a data groupare assigned to SCCC block.

The SCCC_outer_code_mode_A field corresponds to the SCCC outer code modefor Region A within a data group.

The SCCC_outer_code_mode_B field corresponds to the SCCC outer code modefor Region B within the data group.

The SCCC_outer_code_mode_C field corresponds be the SCCC outer code modefor Region C within the data group.

The SCCC_outer_code_mode_D field corresponds to the SCCC outer code modefor Region D within the data group.

The FIC_version field represents a version of FIC data.

The Parade_continuity_counter field counter may increase from 0 to 15and then repeat its cycle. This counter shall increment by 1 every(PRC+1) DATA frames. For example, as shown in Table 12, PRC=011 (decimal3) implies that Parade_continuity_counter increases every fourth DATAframe.

The TNoG field may be identical for all sub-frames in an DATA Frame.

The tpc_protocol_version field is a 5-bit unsigned integer field thatrepresents the version of the structure of the TPC syntax.

TPC data according to the present invention may be extended such that itincludes mobile service data of the E region. In this case, a version ofthe TPC syntax structure indicated by a ‘tpc_protocol_version’ field maybe changed to another version.

TPC data is information for signaling. In the case where the E region isallocated to a transmission area of mobile service data in the datagroup, the TPC data may further include associated informationindicating the above case. One embodiment of the present inventionassumes that scalable mode information is contained in TPC data. Thatis, scalable information indicating an M value from among information of(118+M) mobile service data packets is contained in the TPC data, suchthat the reception end can receive information about the data groupstructure. For example, if the scalable mode is set to ‘000’, M may be‘11’. If the scalable mode is set to ‘001’, M may be ‘20’. If thescalable mode is set to ‘010’, M may be ‘29’. If the scalable mode isset to ‘011’, M may be ‘38’. If the scalable mode is set to ‘111’, M maybe ‘38’ in all data groups transmitted during 16 slot in a sub frame.

In accordance with still another embodiment of the present invention,scalable mode information contained in TPC data may be classified intoscalable mode information of a current frame and scalable modeinformation of the next frame. That is, TPC data contained in thecurrent frame provides the possibility of estimating data to be receivedin the reception end through the next frame's scalable mode information,such that the receiver acting as the reception end can stably receivedata.

However, the information included in the TPC data presented herein ismerely exemplary. And, since the adding or deleting of informationincluded in the TPC may be easily adjusted and modified by one skilledin the art, the present invention will, therefore, not be limited to theexamples set forth herein.

FIG. 19 is a flowchart illustrating a method for processing andtransmitting digital broadcast data according to an embodiment of thepresent invention.

Referring to FIG. 19, a scalable number of mobile service data packetsand a scalable number of main service data packets are multiplexed atstep S1910. For example, mobile service data is contained in (118+M)number of data packets, and main service data is contained in‘156−(118+M)’ number of data packets, where M is 38 or less.

Data contained in the multiplexed data packets is interleaved at stepS1920.

After the interleaving, a data group of up to 156 packets is dispersedthrough 208 data segments.

Interleaved main data and mobile service data contained belonging to onedata group are transmitted in units of a slot at step S1930. In thiscase, mobile service data belonging to the transmitted data group istransmitted through each of 5 regions constructing the data group. Inone embodiment, one data group may be divided into A, B, C, D and Eregions. In this case, the E region may include a scalable number ofmobile service data packets.

The transmitted data group may have any one of data groups shown inFIGS. 3, 4, and 8 to 17. In addition, although the transmitted datagroup is not perfectly identical to each of data groups shown in FIGS.3, 4, and 8 to 17, it may also have a format of any data group havingtechnical characteristics of the present invention.

In accordance with the transmission scheme, data transmitted through theA and B regions and the other data transmitted through the C, D and Eregions may be encoded using different FEC codes, such that the two datamay have different Quality of Services (QoSs). In addition, theFEC-coded data may also be transmitted through all of the A, B, C, D andE regions.

In accordance with the present invention, signaling bytes are insertedinto the A and B regions.

In accordance with the present invention, a fixed number of known datasequences may be inserted into the A and B regions, and a plurality ofknown data may be inserted into the C, D, or E region according to ascalable mode.

FIG. 20 is a flowchart illustrating a method for processing andtransmitting digital broadcast data in such a manner that known data isinserted into regions C and D of a data group, when additional mobileservice data is contained in the data group, according to an embodimentof the present invention.

Provided that a fixed number of mobile service data packets (e.g., 118mobile service data packets) and a scalable number of mobile servicedata packets (e.g., M mobile service data packets) are contained in adata group, if at least one of the scalable mobile service data packetsis contained in the data packet, additional known data may be insertedinto the C and D regions at steps S2010 and S2020.

The fixed and scalable number of mobile service data packets, each ofwhich includes known data in the C and D regions, and the scalablenumber of main service data packets are multiplexed at step S2030.

Data contained in the multiplexed data packet is interleaved at stepS2040.

Interleaved main data and mobile service data contained belonging to onedata group are transmitted in units of a slot at step S2050. In thiscase, mobile service data belonging to the transmitted data group istransmitted through each of 5 regions constructing the data group. Inone embodiment, one data group may be divided into A, B, C, D and Eregions. In this case, the E region may include a scalable number ofmobile service data packets. Known data is inserted into the C and Dregions, such that the reception performance of mobile service datatransmitted through the C and D regions can be increased.

FIG. 21 is a flowchart illustrating a method for processing andtransmitting digital broadcast data in such a manner that mobile servicedata is inserted into a data area for an RS parity and an MPEG headerthat are contained in a data group, when the data group includes 156mobile service data packets, according to an embodiment of the presentinvention.

In the case where one slot is used for transmission of only mobileservice data, no main service data is contained in this slot, such thatthe data group need not have an RS parity and an MPEG header forbackward compatibility.

Therefore, it is determined whether the number of mobile service datapackets contained in the data group is 156 (i.e., M=38) or not at stepS2110.

Assuming that (118+38) mobile service data packets (i.e., 156 mobileservice data packets) are contained in the data group, mobile servicedata is allocated to the data region reserved for the RS parity and theMPEG header that belong to the data group at step S2120.

The data area reserved for the RS parity and the MPEG header multiplexesmobile service data packets of the data group allocated for mobileservice data at step S2130.

Data contained in the multiplexed data packets is interleaved at stepS2140.

Interleaved main data and mobile service data contained belonging to onedata group are transmitted in units of a slot at step S2150. In thiscase, mobile service data belonging to the transmitted data group istransmitted through each of 5 regions constructing the data group. Inone embodiment, one data group may be divided into A, B, C, D and Eregions. In this case, the E region may include a scalable number ofmobile service data packets.

The data area reserved for the RS parity and the MPEG header isallocated to an area for mobile service data, such that much more mobileservice data can be transmitted within one data group.

Receiving System

FIG. 22 illustrates a block view showing the structure of a receivingsystem according to an embodiment of the present invention. Thereceiving system of FIG. 22 includes an antenna 1300, a channelsynchronizer 1301, a channel equalizer 1302, a channel decoder 1303, anRS frame decoder 1304, an M/H TP interface block 1305, a signalingdecoder 1306, an operation controller 1307, an FIC processor 1308, acommon IP protocol stack 1309, an interaction channel unit 1310, an A/Vprocessor 1311, a service signaling channel (SCC) processor 1312, afirst storage unit 1313, a service guide (SG) processor 1314, and asecond storage unit 1315. The receiving system may further include arich media environment (RME) processor 1316, a service protection (SP)processor 1317, and a non-real time (NRT) processor 1318. Also, thereceiving system may further include a main service data processingunit. Herein, the main service data processing unit may include a datadeinterleaver, an RS decoder, and a data derandomizer

According to an embodiment of the present invention, the first storageunit 1313 corresponds to a service map database (DB), and the secondstorage unit 1315 corresponds to a service guide database (DB).

The channel synchronizer 1301 includes a tuner and a demodulator. Thetuner tunes a frequency of a specific channel through the antenna 1300,so as to down-convert the tuned frequency to an intermediate frequency(IF) signal, thereby outputting the converted IF signal to thedemodulator. Herein, the signal being outputted from the tunercorresponds to a passband digital IF signal.

The demodulator included in the channel synchronizer 1301 uses knowndata sequences included in a data group and transmitted from thetransmitting system, so as to perform carrier wave recovery and timingrecovery, thereby converting the inputted pass band digital signal to abaseband digital signal.

For example, among the known data sequences, the channel equalizer 1302uses a 1st known data sequence, and 3rd to 6th known data sequences tocompensate the distortion in a received signal caused by multi path or aDoppler effect. At this point, the channel equalizer 1302 may enhancethe equalizing performance by being fed-back with the output of thechannel decoder 1303.

The signaling decoder 1306 extracts signaling data (e.g., TPC data andFIC data) from the received signal and decodes the extracted signaldata. The decoded TPC data are outputted to the operation controller1307, and the decoded FIC data are outputted to the FIC processor 1308.According to an embodiment of the present invention, the signalingdecoder 1306 performs signaling decoding as an inverse process of thesignaling encoder, so as to extract the TPC data and the FIC data fromthe received signal. For example, the signaling decoder 1306 performs aparallel concatenated convolution code (PCCC) type regressive turbodecoding process on the data corresponding to the signaling informationregion within the inputted data. Then, the signaling decoder 1306derandomizes the turbo-decoded signaling data, thereby separating theTPC data and the FIC data from the derandomized signaling data.Additionally, the signaling decoder 1306 performs RS-decoding on theseparated TPC data as an inverse process of the transmitting system,thereby outputting the RS-decoded TPC data to the operation controller1307.

Herein, the TPC data may include RS frame information, SCCC information,M/H frame information, and so on. The RS frame information may includeRS frame mode information and RS code mode information. The SCCCinformation may include SCCC block mode information and SCCC outer codemode information. The M/H frame information may include indexinformation. And, the TPC data may include subframe count information,slot count information, parade_id information, SGN information, NOGinformation, and so on.

Thereafter, the signaling decoder 1306 performs deinterleaving on theseparated FIC data in subframe units and performs RS decoding on thedeinterleaved data as an inverse process of the transmitting system,thereby outputting the RS-decoded data to the FIC processor 1308. Thetransmission unit of the FIC data being deinterleaved and RS-decoded bythe signaling decoder 1306 and outputted to the FIC processor 1308corresponds to FIC segments.

The channel decoder 1303, which is also referred to as a block decoder,performs forward error correction in order to recover meaningful data(e.g., mobile service data) from the received signal. According to anembodiment of the present invention, in order to do so, the channeldecoder 1303 uses SCCC-related information (e.g., SCCC blockinformation, SCCC outer code mode information, and so on) included inthe signaling data. According to the embodiment of the presentinvention, if the data being channel-equalized and inputted from thechannel equalizer 1302 correspond to data processed with both serialconcatenated convolution code (SCCC) type block-encoding andtrellis-encoding (i.e., data within the RS frame, signaling data) by thetransmitting system, the channel decoder 1303 performs trellis-decodingand block-decoding on the corresponding data as an inverse process ofthe transmitting system. Alternatively, if the data beingchannel-equalized and inputted from the channel equalizer 1302correspond to data processed only with trellis-encoding and not withblock-encoding (i.e., main service data), the channel decoder 1303performs only trellis-decoding on the corresponding data.

By performing RS-CRC decoding on the received data, the RS frame decoder1304 recovers the RS frame. More specifically, the RS frame decoder 1304performs forward error correction in order to recover the RS frame. Inorder to do so, according to an embodiment of the present invention, theRS frame decoder 1304 uses RS-associated information (e.g., RS codemode) among the signaling data.

The M/H TP interface block 1305 extracts M/H TP packets from the RSframe, so as to recover the IP datagram, thereby outputting therecovered IP datagram to the common IP protocol stack 1309. Herein, theM/H TP packets encapsulate the IP datagram. More specifically, theheader of each M/H TP packet is analyzed so as to recover the IPdatagram from the payload of the corresponding M/H TP packet.

The operation controller 1307 uses the decoded TPC data structure so asto control the operations of various baseband processes. Morespecifically, the operation controller 1307 receives the TPC data anddelivers information, such as M/H frame timing information, informationon whether or not a data group exists in a selected parade, positioninformation of known data within the data group, and power controlinformation, to block requiring the respective information.

The FIC processor 1308 collects (or gathers) FIC segments to recover anFIC chunk and stores the recovered FIC chunk in the first storage unit1313. The FIC chunk includes signaling information required in anensemble selection process and a mobile (or M/H) service scanningprocess.

The service signaling channel processor 1312 extracts service signalingchannel table sections from the designated IP multicast streams andstores the extracted service signaling channel table sections in thefirst storage unit 1313. The service signaling channel includes IP levelsignaling information, which is required for M/H service selection andscanning processes. Herein, the service signaling channel according tothe present invention transmits at least one of an SMT, a GAT, an RRT, aCIT, and an SLT. At this point, according to embodiment of the presentinvention, the access information of the IP datagram transmitting theservice signaling channel corresponds to a well-known destination IPaddress and a well-known destination UDP port number. Therefore, theservice signaling channel processor 1312 has a well-known destination IPaddress and a well-known destination UDP port number, thereby beingcapable of extracting the IP stream transmitting the service signalingchannel, i.e., service signaling data. Then, at least one of the SMT,the GAT, the RRT, the CIT, and the SLT extracted from the servicesignaling data is recovered and stored in the first storage unit 1313.For example, the first storage unit 1313 stores a service map configuredfrom signaling information collected the FIC processor 1308 and theservice signaling channel processor 1312.

The A/V processor 1311 receives the IP datagram from the common IPprotocol stack 1309. Then, the A/V processor 1311 separates the audiodata and the video data from the received IP datagram and decoded eachof the audio data and the video data with a respective decodingalgorithm, thereby displaying the decoded data to the screen. Forexample, at least one of an AC-3 decoding algorithm, an MPEG 2 audiodecoding algorithm, an MPEG 4 audio decoding algorithm, an AAC decodingalgorithm, an AAC+ decoding algorithm, an HE AAC decoding algorithm, anAAC SBR decoding algorithm, an MPEG surround decoding algorithm, and aBSAC decoding algorithm may be applied be used as the audio decodingalgorithm, and at least one of an MPEG 2 video decoding algorithm, anMPEG 4 video decoding algorithm, an H.264 decoding algorithm, an SVCdecoding algorithm, and a VC-1 decoding algorithm may be applied as thevideo decoding algorithm.

The SG processor 1314 recovers announcement data and stores therecovered announcement data to the second storage unit 1315, therebyproviding a service guide to the viewer.

The interaction (or return) channel unit 1310 provides an uplink fromthe receiving system through the common IP protocol stack 1309. At thispoint, the interaction channel should be IP-compatible.

The RME processor 1316 receives an M/H broadcast program or RME datathrough the common IP protocol stack 1309, the RME data being deliveredthrough the interaction channel. Then, the received M/H broadcastprogram or RME data are recovered and then processed.

The SP processor 1317 recovers and processes data associated withservice protection, which are received through the common IP protocolstack 1309. Then, the SP processor 1317 provides protection to the M/Hservice depending upon the subscription state of the viewer (or user).

The NRT processor 1318 recovers and processes non-real time data, suchas file application.

Channel Synchronizer

FIG. 23 illustrates a detailed block view of a demodulator included inthe channel synchronizer 1301 according to an embodiment of the presentinvention.

The channel synchronizer 1301 of FIG. 23 may include a phase splitter1501, a first multiplier 1502, a resampler 1503, a Matched Filter 1504,a Timing Recovery block 1505, a Group Position Synchronization & InitialFrequency offset estimator 1506, a Carrier Recovery block 1507, and a DCremover 1508. According to an embodiment of the present invention, ananalog-to-digital converter (not shown) converting a passband analogsignal to a passband digital signal may be provided at the front end ofthe phase splitter 1501. Also, according to an embodiment of the presentinvention, an automatic gain control (AGC) is performed before carrierrecovery and timing recovery.

The carrier recovery block 1507 includes a delay unit 1601, a secondmultiplier 1603, a carrier frequency offset detector 1604, a multiplexer1605, a loop filter 1606, and Numerically Controlled Oscillator (NCO)1607.

Also referring to FIG. 23, the phase splitter 1501 receives a pass banddigital signal and splits the received signal into a pass band digitalsignal of a real number element and a pass band digital signal of animaginary number element both having a phase of 90 degrees between oneanother. In other words, the pass band digital signal is split intocomplex signals. The split portions of the pass band digital signal arethen outputted to the first multiplier 1502. Herein, the real numbersignal outputted from the phase splitter 1501 will be referred to as anT signal, and the imaginary number signal outputted from the phasesplitter 1501 will be referred to as a ‘Q’ signal, for simplicity of thedescription of the present invention.

It is assumed that the signal being outputted from the tuner of thepresent invention is an intermediate frequency (IF) of 44 MHz. Accordingto another embodiment of the present invention, the signal beingoutputted from the tuner may also be a zero IF signal (i.e., complexbaseband signal). In this case, the zero IF signal is inputted to thefirst multiplier 1502 bypassing the phase splitter 1501.

The first multiplier 1502 multiplies the I and Q pass band digitalsignals, which are outputted from the phase splitter 1501, by a digitalcomplex signal outputted from the NCO 1607 of the carrier recovery block1507, thereby down-converting the I and Q passband digital signals tobaseband digital complex signals. At this point, by multiplying thecarrier frequency offset being outputted from the carrier recovery block1507 by the output of the phase splitter 1501, the carrier frequencyoffset included in the output signal of the phase splitter 1501 iscompensated. Thereafter, the baseband digital signals of the firstmultiplier 1502 are inputted to the resampler 1503.

The resampler 1503 multiplies the signals outputted from the firstmultiplier 1502 by a sampling clock provided by the timing recoveryblock 1505, so as to compensate symbol timing errors, thereby outputtingthe compensated signals to the matched filter 1504 and the timingrecovery block 1505.

The matched filter 1504 performs matched filtering on the output signalsof the resampler 1503, thereby outputting the signals processed withmatched filtering to the Group Position Synchronization & InitialFrequency offset estimator 1506, the Carrier Recovery block 1507, andthe DC remover 1508.

The Group Position Synchronization & Initial Frequency offset estimator1506 detects the place (or position) of the known data sequences thatare included in the data group and received. Simultaneously, the GroupPosition Synchronization & Initial Frequency offset estimator 1506estimates an initial frequency offset during the known data detectionprocess. In this case, the carrier recovery block 1507 may use the knowndata sequence position information and initial frequency offset value toestimate the carrier frequency offset with more accuracy, therebyperforming compensation. Also, the Group Position Synchronization &Initial Frequency offset estimator 1506 performs group positionsynchronization. More specifically, the Group Position Synchronization &Initial Frequency offset estimator 1506 extracts the starting positionof each data group.

For example, the Group Position Synchronization & Initial Frequencyoffset estimator 1506 detects the position (or place) of the known datasequence included in the data group. Then, the Group PositionSynchronization & Initial Frequency offset estimator 1506 outputs thedetected known sequence position indicating signal to the multiplexer1604 and the channel equalizer 1302 of the carrier recovery block 1507.Furthermore, the Group Position Synchronization & Initial Frequencyoffset estimator 1506 estimates the initial frequency offset by usingthe second known data sequence within the data group, which is thenoutputted to the loop filter 1606 of the carrier recovery block 1507.

The timing recovery block 1505 receives the output of the resampler 1503so as to detect the timing error. Then, the timing recovery block 1505outputs a sampling clock being in proportion with the detected timingerror to the resampler 1503, thereby controlling the sampling of theresampler 1503.

The DC remover 1508 removes a pilot tone signal (i.e., DC signal), whichhas been inserted by the transmitting system, from the matched-filteredsignal outputted from the matched filter 1504. Thereafter, the DCremover 1508 outputs the processed signal to the channel equalizer 1302.

Known Data Sequence Position and Initial Frequency Offset Estimation

According to an embodiment of the present invention, among the knowndata sequences included in the data group, a correlation of repeatedknown data patterns of a second known data sequence is used to detectthe position of a known data sequence within the data group and toestimate an initial frequency offset. Particularly, according to anembodiment of the present invention, a partial correlation is used todetect the position of a known data sequence and to estimate an initialfrequency offset.

The initial frequency offset includes a rough frequency offset and afiner frequency offset. More specifically, when acquiring carrierfrequency acquisition, at first a rough frequency offset is used toreduce a frequency pull-in range, and, then, a finer frequency offset isused to reduce the frequency pull-in range once again.

In the present invention, the second known data sequence within the datagroup is configured of a first 528 symbol sequence and a second 528symbol sequence each having the same pattern. More specifically, the 528pattern is repeated after a data segment synchronization signal of 4symbols.

In the description of the present invention, the second known datasequence will be referred to as an acquisition training sequence.

The Group Position Synchronization & Initial Frequency offset estimator(or known sequence detector) 1506 uses the repeated known data patternof the second known data sequence, so as to perform group positionsynchronization and initial frequency offset estimation. For example, acorrelation between the received signal and a second known datasequence, which is pre-known based upon an agreement between thetransmitting system and the receiving system, and by finding a maximumcorrelation value, the group position synchronization may be performed.However, if a frequency offset exists in the received signal, thereliability of a general correlation method, wherein an entire secondknown data sequence is used to obtain one correlation value, may bedegraded. More specifically, as the length of a known data pattern forcorrelation becomes longer, the possibility of the reliability of acorrelation peak value being degraded may increase.

Therefore, according to an embodiment of the present invention, apartial correlation method is used to detect a highly reliablecorrelation peak value, even when a large frequency offset exists. Morespecifically, by using the partial correlation method, noise may bereduced.

AS described above, in the present invention, by obtaining a partialcorrelation by dividing (or segmenting) the second known data sequenceinto two or more parts, a peak value among the correlation value foreach part may be obtained. Then, all of the peak correlation values foreach part are added so as to calculate the average (or mean) value.Accordingly, the noise included in the received signal is cancelled,thereby reducing the noise.

In order to do so, the second known data sequence is segmented (ordivided) into multiple parts, and a correlation between the known datasequence of each part (i.e., reference known data sequence of acorresponding part generated from the receiving system) and thereceiving signal is calculated (or obtained) for each part. Thereafter,the partial correlation values obtained for each part are all added. Atthis point, each of the correlation values obtained for each partcorresponds to a squared value (i.e., a magnitude square) having nophase information.

(a) of FIG. 24 shows an example of each part being configured of Nnumber of symbols, when the second known data symbol sequence is divided(or segmented) into L number of parts. More specifically, in (a) of FIG.24, L represents a number of parts being segmented from the second knowndata symbol sequence, and N represents the length of each part. Also, *signifies a complex conjugate. In other words, a second known datasequence is divided into L number of parts each having the length of Nsymbols. Thereafter, the correlation with the received signal isobtained for each part.

(b) of FIG. 24 illustrates a conceptual view of a partial correlatoraccording to an embodiment of the present invention. Herein, the partialcorrelator consists of a multiplier 1701 shifting known data sequencesof each corresponding part to the received signal so as to performcomplex conjugate multiplication, a first accumulator 1702 accumulatingthe output of the multiplier 1701 for a period of N symbols, a squarer1703 calculating a squared value of the output of the first accumulator1702, and a second accumulator 1704 accumulating the output of thesquarer 1703 for a predetermined period of time, thereby calculating anaverage (or mean) correlation value.

More specifically, the multiplier 1701 shifts the known data sequence ofa corresponding part in accordance with the received signal so as toperform complex conjugate multiplication, thereby outputting themultiplied values to the first accumulator 1702. The first accumulator1702 accumulates the output of the multiplier 1701 for a period of Nsymbols, thereby outputting the accumulated value to the squarer 1703.The output of the first accumulator 1702 corresponds to correlationvalues each having a phase and size. The squarer 1703 calculates thesquared value of the output of the first accumulator 1702, therebyobtaining the size of the correlation value. The second accumulator 1704accumulates the outputs of the squarer 1703 during L sections. Then, thesecond accumulator 1704 divides the accumulated value by L, so as tooutput the average correlation value of the corresponding part. Equation1 shown below corresponds to (b) of FIG. 24 recapitulated in the form ofan equation.

$\begin{matrix}{{y\lbrack k\rbrack} = {{1/L}{\sum\limits_{i = 0}^{L}\; {{z\left\lbrack {k + {N\; }} \right\rbrack}}^{2}}}} & {{Equation}\mspace{14mu} 1}\end{matrix}$

More specifically, by finding a maximum correlation value during a datagroup period, the Group Position Synchronization & Initial Frequencyoffset estimator 1506 decides a group synchronization position and aknown data sequence position. Also, by suing the partial correlationmethod, the known sequence detector estimates the initial frequencyoffset.

FIG. 25 illustrates an example of estimating a rough initial frequencyoffset by obtaining partial correlation by dividing (or segmenting) asecond known data sequence into 8 parts. When applying this example toFIG. 24, L is equal to 8, and N is equal to 132 symbols. In case of FIG.25, 8 partial correlators are required to be provided, and 8 peakcorrelation values may be obtained accordingly.

In this case, at a maximum correlation position, the Group PositionSynchronization & Initial Frequency offset estimator 1506 calculates aphase difference between the correlation values of each of thesuccessive parts. Then, the Group Position Synchronization & InitialFrequency offset estimator 1506 uses an adder 1801 to add all of thephase differences for each part, thereby outputting an average phasedifference Δθ. Subsequently, by using the average phase difference Δθand the length (N) of each part, the known sequence detector maycalculate ω0 as shown in Equation 2 below.

$\begin{matrix}{\omega_{0} = \frac{\Delta \; \theta}{N}} & {{Equation}\mspace{14mu} 2}\end{matrix}$

Herein,

-   -   ω₀=2πf0    -   f0: Normalized frequency offset    -   Δθ: Average phase difference)    -   N: Length of each part

According to an embodiment of the present invention, in Equation 2, N isequal to 132.

At this point, the rough (or coarse) frequency offset f0 obtained fromω0 by applying Equation 2 provides a frequency pull-in range of ±80 kHz.A trade-off exists between the frequency pull-in range and a variance ofan estimated error associated to a length of the part. Morespecifically, if the length of a known data pattern for the correlationis short, the frequency pull-in range increases, and the jitter alsoincreases accordingly. On the other hand, if the length of a known datapattern for the correlation is long, the frequency pull-in rangedecreases, and the jitter also decreases accordingly.

Meanwhile, according to an embodiment of the present invention, a finerfrequency offset is estimated by using repeated patterns of the secondknown data sequence. The second known data sequence is configured of twoparts. More specifically, the second known data sequence is configuredof a part including a first 528 symbol sequence and another partincluding a second 528 symbol sequence. Herein, a data segmentsynchronization signal of 4 symbols exists between the first 528 symbolsequence and the second 528 symbol sequence. This structure allows thefiner frequency offset to be estimated by using a Maximum-likelihoodalgorithm.

FIG. 26 illustrates an example of estimating a finer frequency offset byusing the Maximum-likelihood algorithm according to the presentinvention.

At this point, the received signal al may be indicated as shown inEquation 3 below.

r[k]=x[k] ^(−j2πf) ⁰ ^(T) ^(s) ^(k) +n[k]  Equation 3

Herein,

-   -   x[k]: transmitted signal    -   f0: frequency offset    -   Ts: symbol duration    -   n[k]: noise

In Equation 3, f0 corresponds to the finer frequency offset.

Also, the correlation between the received signals separated by 532(=528+4) symbols may be obtained (or calculated) by using Equation 4below.

$\begin{matrix}\begin{matrix}{{E\left\{ {{r\lbrack k\rbrack}{r^{*}\left\lbrack {k + 532} \right\rbrack}} \right\}} = {E\left\{ \left( {{{x\lbrack k\rbrack}^{{- {j2\pi}}\; f_{0}T_{s}k}} +} \right. \right.}} \\{\left. {n\lbrack k\rbrack} \right)\left( {{{x^{*}\left\lbrack {k + 532} \right\rbrack}^{{j2\pi}\; f_{0}{T_{s}{({k + 532})}}}} +} \right.} \\\left. \left. {n^{*}\left\lbrack {k + 532} \right\rbrack} \right) \right\} \\{= {\sigma_{s}^{2}^{{j2\pi}\; f_{0}{T_{s} \cdot 532}}}}\end{matrix} & {{Equation}\mspace{14mu} 4}\end{matrix}$

More specifically, the partial correlation of the two parts spaced apartby 532 symbols (i.e., each part having the length of 532 symbols) iscalculated and then the average value is calculated. Thereafter, afterapplying an argument, the finer frequency offset can be obtained. Thefiner frequency offset obtained by applying the Equation 4 provides afrequency pull-in range of ±10 kHz.

In the description of the present invention, the rough frequency offsetand the finer frequency offset will be collectively referred to as aninitial frequency offset. The initial frequency offset is outputted tothe loop filter 1606 of the carrier recovery block 1507.

Meanwhile, according to an embodiment of the present invention, thecarrier recovery block 1507 uses a carrier frequency tracking loop, asshown in FIG. 23.

The carrier recovery block 1507 loads an initial frequency offsetestimated from the Group Position Synchronization & Initial Frequencyoffset estimator 1506. Then, the carrier recovery block 1507 tracks theremaining carrier frequency offset.

More specifically, the carrier recovery block 107 uses aMaximum-likelihood to calculate the correlation of the receivedsuccessive known data sequences, thereby estimating a carrier frequencyoffset (or error) using the same method that is used for the initialfrequency offset estimation.

In order to do so, the delay unit 1601 of the carrier recovery block1507 receives the data outputted from the matched filter 1504 in symbolunits so as to perform a K symbol delay. Thereafter, the delay unit 1601outputs the delayed data to the second multiplier 1603.

Also, the output data of the matched filter 1504 is conjugated by theconjugator 1602. Then, the conjugated data are inputted to the secondmultiplier 1603.

The second multiplier 1603 calculates the correlation value between theknown data sequence delayed by K symbols by the delay unit 1601 and theknown data sequence conjugated by the conjugator 1602. Thereafter, thesecond multiplier 1603 outputs the calculated correlation value to thecarrier frequency offset detector 1604.

Herein, according to an embodiment of the present invention, K symbolsis equal to 13312 symbols (=832*16 symbols). This is because the firstknown data sequence, and the third to sixth known data sequences areinserted and received at intervals of 16 segments, and also because onesegment is configured of 832 symbols.

According to the embodiment of the present invention, the correlationvalue between the known data sequences spaced apart at an interval of13312 symbols may be calculated by applying Equation 5 shown below.

$\begin{matrix}\begin{matrix}{{E\left\{ {{r\lbrack k\rbrack}{r^{*}\left\lbrack {k + 13312} \right\rbrack}} \right\}} = {E\left\{ \left( {{{x\lbrack k\rbrack}^{{- {j2\pi}}\; f_{0}T_{s}k}} +} \right. \right.}} \\{\left. {n\lbrack k\rbrack} \right)\left( {{{x^{*}\left\lbrack {k + 13312} \right\rbrack}^{{j2\pi}\; f_{0}{T_{s}{({k + 13312})}}}} +} \right.} \\\left. \left. {n^{*}\left\lbrack {k + 13312} \right\rbrack} \right) \right\} \\{= {\sigma_{s}^{2}^{{j2\pi}\; f_{0}{T_{s} \cdot 13312}}}}\end{matrix} & {{Equation}\mspace{14mu} 5}\end{matrix}$

Where, σ_(s) ²: E{|x[k]|²}

Herein,

-   -   x[k]: transmitted signal    -   i. f0: carrier frequency offset    -   ii. Ts: symbol duration    -   iii. n[k]: noise

In Equation 5, f0 corresponds to a carrier frequency offset fortracking.

The carrier frequency offset detector 1604 extracts a carrier frequencyoffset from the correlation value outputted from the second multiplier1603, as shown in Equation 11. Then, the extracted carrier frequencyoffset is outputted to the multiplexer 1605.

In accordance with the Known Sequence Position Indicating Signal fromthe Group Position Synchronization & Initial Frequency offset estimator1506, the multiplexer 1605 selects an output of the carrier frequencyoffset detector 1604 or ‘0’, which is then outputted as the finalcarrier frequency offset value.

More specifically, by using Known Sequence Position Indicating Signal,the validity of the carrier frequency offset value being outputted fromthe carrier frequency offset detector 1604 can be known. If the carrierfrequency offset value is valid, the multiplexer 1605 selects the outputof the carrier frequency offset detector 1604. And, if the carrierfrequency offset value is not valid, the multiplexer 1605 selects ‘0’.Then, the multiplexer 1605 outputs the selection to the loop filter1606.

The loop filter 1606 adds the output of the multiplexer 1605 to theestimated initial frequency offset, so as to perform basebandpass-filtering. Thereafter, the filtered data are outputted to the NCO1607.

The NCO 1607 generates a complex signal corresponding to a basebandpass-filtered carrier frequency offset, thereby outputting the generatedcomplex signal to the first multiplier 1502.

Meanwhile, according to an embodiment of the present invention, byturning the power on only in particular slots, i.e., slots having thedata groups of a parade allocated thereto, wherein the parade includes amobile service requested to be received, the channel synchronizer 1301may reduce power consumption in the receiving system. For this, thereceiving system may further include a power controller (not shown),which controls the power supply of the demodulator.

Channel Equalizer

The data demodulated by the channel synchronizer 1301 using the knowndata are inputted to the channel equalizer 1302. Also, the demodulateddata may be inputted to the known sequence detector 1506.

At this point, a data group that is inputted for the equalizationprocess may be divided into region A to region D, as shown in FIG. 4.More specifically, according to the embodiment of the present invention,region A includes M/H block B4 to M/H block B7, region B includes M/Hblock B3 and M/H block B8, region C includes M/H block B2 and M/H blockB9, and region D includes M/H block B1 and M/H block B10. In otherwords, one data group is divided into M/H blocks from B1 to B10, eachM/H block having the length of 16 segments. Also, a long trainingsequence (i.e., known data sequence) is inserted at the starting portionof the M/H blocks B4 to B8. Furthermore, two data groups may beallocated (or assigned) to one VSB field. In this case, fieldsynchronization data are positioned in the 37^(th) segment of one of thetwo data groups.

The present invention may use known data, which have position andcontent information based upon an agreement between the transmittingsystem and the receiving system, and/or field synchronization data forthe channel equalization process.

The channel equalizer 1302 may perform channel equalization using aplurality of methods. According to the present invention, the channelequalizer 2003 uses known data and/or field synchronization data, so asto estimate a channel impulse response (CIR), thereby performing channelequalization.

Most particularly, an example of estimating the CIR in accordance witheach region within the data group, which is hierarchically divided andtransmitted from the transmitting system, and applying each CIRdifferently will also be described herein.

At this point, a data group can be assigned and transmitted a maximumthe number of 4 in a VSB frame in the transmitting system. In this case,all data group do not include field synchronization data. In the presentinvention, the data group including the field synchronization dataperforms channel-equalization using the field synchronization data andknown data. And the data group not including the field synchronizationdata performs channel-equalization using the known data.

For example, the data of the M/H block B3 including the fieldsynchronization data performs channel-equalization using the CIRcalculated from the field synchronization data area and the CIRcalculated from the first known data area. Also, the data of the M/Hblocks B1 and B2 performs channel-equalization using the CIR calculatedfrom the field synchronization data area and the CIR calculated from thefirst known data area. Meanwhile, the data of the M/H blocks B1 to B3not including the field synchronization data performschannel-equalization using CIRS calculated from the first known dataarea and the third known data area.

As described above, the present invention uses the CIR estimated fromthe known data region in order to perform channel equalization on datawithin the data group. At this point, each of the estimated CIRs may bedirectly used in accordance with the characteristics of each regionwithin the data group. Alternatively, a plurality of the estimated CIRsmay also be either interpolated or extrapolated so as to create a newCIR, which is then used for the channel equalization process.

Herein, when a value F(Q) of a function F(x) at a particular point Q anda value F(S) of the function F(x) at another particular point S areknown, interpolation refers to estimating a function value of a pointwithin the section between points Q and S. Linear interpolationcorresponds to the simplest form among a wide range of interpolationoperations. FIG. 27 illustrates an example of linear interpolation.

More specifically, in a random function F(x), when given the values F(Q)and F(S) each from points x=Q and x=S, respectively, the approximatevalue F(P) of the F(x) function at point x=P may be estimated by usingEquation 7 below. In other words, since the values of F(Q) and F(S)respective to each point x=Q and x=S are known (or given), a straightline passing through the two points may be calculated so as to obtainthe approximate value F(P) of the corresponding function value at pointP.

At this point, the straight line passing through points (Q,F(Q)) and(S,F(S)) may be obtained by using Equation 6 below.

$\begin{matrix}{{\hat{F}(x)} = {{\frac{{F(S)} - {F(Q)}}{S - Q}\left( {x - Q} \right)} + {F(Q)}}} & {{Equation}\mspace{14mu} 6}\end{matrix}$

Accordingly, Equation 7 below shows the process of substituting p for xin Equation 6, so as to calculate the approximate value F(P) of thefunction value at point P.

$\begin{matrix}{{{\hat{F}(P)} = {{\frac{{F(S)} - {F(Q)}}{S - Q}\left( {P - Q} \right)} + {F(Q)}}}{{\hat{F}(P)} = {{\frac{S - P}{S - Q}{F(Q)}} + {\frac{P - Q}{S - Q}{F(S)}}}}} & {{Equation}\mspace{14mu} 7}\end{matrix}$

The linear interpolation method of Equation 7 is merely the simplestexample of many other linear interpolation methods. Therefore, since anyother linear interpolation method may be used, the present inventionwill not be limited only to the examples given herein.

Alternatively, when a value F(Q) of a function F(x) at a particularpoint Q and a value F(S) of the function F(x) at another particularpoint S are known (or given), extrapolation refers to estimating afunction value of a point outside of the section between points Q and S.Herein, the simplest form of extrapolation corresponds to linearextrapolation.

FIG. 28 illustrates an example of linear extrapolation. As describedabove, for linear extrapolation as well as linear interpolation, in arandom function F(x), when given the values F(Q) and F(S) each frompoints x=Q and x=S, respectively, the approximate value {circumflex over(F)}(P) of the corresponding function value at point P may be obtainedby calculating a straight line passing through the two points.

Herein, linear extrapolation is the simplest form among a wide range ofextrapolation operations. Similarly, the linear extrapolation describedherein is merely exemplary among a wide range of possible extrapolationmethods. And, therefore, the present invention is not limited only tothe examples set forth herein

FIG. 29 illustrates a block diagram of a channel equalizer according toan embodiment of the present invention.

Referring to FIG. 29, the channel equalizer includes a first frequencydomain converter 4100, a channel estimator 4110, a second frequencydomain converter 4121, a coefficient calculator 4122, a distortioncompensator 4130, and a time domain converter 4140.

Herein, the channel equalizer may further include a remaining carrierphase error remover, a noise canceller (NC), and a decision unit.

The first frequency domain converter 4100 includes an overlap unit 4101overlapping inputted data, and a fast fourier transform (FFT) unit 4102converting the data outputted from the overlap unit 4101 to frequencydomain data.

The channel estimator 4110 includes a CIR estimator 4111, a firstcleaner 4112, a multiflexer 4113, a CIR calculator 4114, a secondcleaner 4115 and a zero-padding unit 4116.

Herein, the channel estimator 4110 may further include a phasecompensator compensating a phase of the CIR which estimated in the CIRestimator 4111.

The second frequency domain converter 4121 includes a fast fouriertransform (FFT) unit converting the CIR being outputted from the channelestimator 4110 to frequency domain CIR.

The time domain converter 4140 includes an IFFT unit 4141 converting thedata having the distortion compensated by the distortion compensator4130 to time domain data, and a save unit 4142 extracting only validdata from the data outputted from the IFFT unit 4141. The output datafrom the save unit 4142 corresponds to the channel-equalized data.

If the remaining carrier phase error remover is connected to an outputterminal of the time domain converter 4140, the remaining carrier phaseerror remover estimates the remaining carrier phase error included inthe channel-equalized data, thereby removing the estimated error.

If the noise remover is connected to an output terminal of the timedomain converter 4140, the noise remover estimates noise included in thechannel-equalized data, thereby removing the estimated noise.

More specifically, the receiving data demodulated in FIG. 29 areoverlapped by the overlap unit 4101 of the first frequency domainconverter 4100 at a pre-determined overlapping ratio, which are thenoutputted to the FFT unit 4102. The FFT unit 4102 converts theoverlapped time domain data to overlapped frequency domain data throughby processing the data with FFT. Then, the converted data are outputtedto the distortion compensator 4130.

The distortion compensator 4130 performs a complex number multiplicationon the overlapped frequency domain data outputted from the FFT unit 4102included in the first frequency domain converter 4100 and theequalization coefficient calculated from the coefficient calculator4122, thereby compensating the channel distortion of the overlapped dataoutputted from the FFT unit 4102. Thereafter, the compensated data areoutputted to the IFFT unit 4141 of the time domain converter 4140. TheIFFT unit 4141 performs IFFT on the overlapped data having the channeldistortion compensated, thereby converting the overlapped data to timedomain data, which are then outputted to the save unit 4142. The saveunit 4142 extracts valid data from the data of the channel-equalized andoverlapped in the time domain, and outputs the extracted valid data.

Meanwhile, the received data are inputted to the overlap unit 4101 ofthe first frequency domain converter 4100 included in the channelequalizer and, at the same time, inputted to the CIR estimator 4111 ofthe channel estimator 4110.

The CIR estimator 4111 uses a training sequence, for example, data beinginputted during the known data section and the known data in order toestimate the CIR. If the data to be channel-equalizing is the datawithin the data group including field synchronization data, the trainingsequence using in the CIR estimator 4111 may become the fieldsynchronization data and known data. Meanwhile, if the data to bechannel-equalizing is the data within the data group not including fieldsynchronization data, the training sequence using in the CIR estimator4111 may become only the known data.

For example, the CIR estimator 4111 uses the data received during aknown data section and reference known data generated from the receivingsystem based upon an agreement between the receiving system and thetransmitting system, so as to estimate a channel impulse response (CIR).In order to do so, the CIR estimator 4111 is provided with a KnownSequence Position Indicating Signal from the Group PositionSynchronization & Initial Frequency offset estimator 1506.

Also, in case of the data group including field synchronization, the CIRestimator 4111 may use the data being received during one fieldsynchronization section and the reference field synchronization data,which generated from the receiving system in accordance with anagreement between the transmitting system and the receiving system, soas to estimate a channel impulse response (CIR_FS). In order to do so,the CIR estimator 4111 may be provided with Field Sync PositionInformation from the Group Position Synchronization & Initial Frequencyoffset estimator 1506. The CIR estimator 4111 may estimate a channelimpulse response (CIR) by using a well-known least square (LS) method.

In the LS method, a cross correlation value p between known data thathave passed through a channel during a known data section and known dataalready known by a receiving end (or receiver) is calculated, and anauto-correlation matrix R of the known data is also calculated.Thereafter, a matrix operation (or calculation) of R⁻¹·p is performed sothat the auto-correlation portion existing in the cross correlationvalue p between the received data and the initial (or original) knowndata can be removed, thereby estimating the CIR of the transmissionchannel.

Also, according to another embodiment of the present invention, the CIRestimator may also perform CIR estimation by using a least mean square(LMS) method. For example, in regions A and B within the data group, theChannel Impulse Response (CIR) is estimated by using the Least Square(LS) method, and, then, channel equalization may be performed.Thereafter, in regions C and D within the data group, the CIR isestimated by using the Least Mean Square (LMS) method, and, then,channel equalization may be performed.

The CIR estimated as described above is outputted to the first cleaner4112 and the multiplexer 4113. The multiplexer 4113 may either selectthe output of the first cleaner 4112 or select the output of the CIRestimator 4111 depending upon whether the CIR operator 4114 performsinterpolation on the estimated CIR, or whether the CIR operator 4114performs extrapolation on the estimated CIR. For example, when the CIRoperator 4114 performs interpolation on the estimated CIR, themultiplexer 4113 selects the output of the CIR estimator 4111. And, whenthe CIR operator 4114 performs extrapolation on the estimated CIR, themultiplexer 4113 selects the output of the first cleaner 4112.

The CIR operator 4114 performs interpolation or extrapolation on theestimated CIR and then outputs the interpolated or extrapolated CIR tothe second cleaner 4115.

More specifically, the CIR estimated from the known data includes achannel element that is to be obtained as well as a jitter elementcaused by noise. Since such jitter element deteriorates the performanceof the equalizer, it preferable that a coefficient calculator 4122removes the jitter element before using the estimated CIR. Therefore,according to the embodiment of the present invention, each of the firstand second cleaners 4113 and 4115 removes a portion of the estimated CIRhaving a power level lower than the predetermined threshold value (i.e.,so that the estimated CIR becomes equal to ‘0’). Herein, this removalprocess will be referred to as a “CIR cleaning” process.

The CIR calculator 4114 performs CIR interpolation by multiplying CIRsestimated from the CIR estimator 4111 by each of coefficients, therebyadding the multiplied values. At this point, some of the noise elementsof the CIR may be added to one another, thereby being cancelled.Therefore, when the CIR calculator 4114 performs CIR interpolation, theoriginal (or initial) CIR having noise elements remaining therein. Inother words, when the CIR calculator 4114 performs CIR interpolation,the estimated CIR bypasses the first cleaner 4113 and is inputted to theCIR calculator 4114. Subsequently, the second cleaner 4115 cleans theCIR interpolated by the CIR interpolator-extrapolator 4114.

Conversely, the CIR calculator 4114 performs CIR extrapolation by usinga difference value between two CIRs, so as to estimate a CIR positionedoutside of the two CIRs. Therefore, in this case, the noise element israther amplified. Accordingly, when the CIR calculator 4114 performs CIRextrapolation, the CIR cleaned by the first cleaner 4113 is used. Morespecifically, when the CIR calculator 4114 performs CIR extrapolation,the extrapolated CIR passes through the second cleaner 4115, therebybeing inputted to the zero-padding unit 4116.

Meanwhile, when a second frequency domain converter (or fast fouriertransform (FFT2)) 4121 converts the CIR, which has been cleaned andoutputted from the second cleaner 4115, to a frequency domain, thelength and of the inputted CIR and the FFT size may not match (or beidentical to one another). In other words, the CIR length may be smallerthan the FFT size. In this case, the zero-padding unit 4116 adds anumber of zeros ‘0’s corresponding to the difference between the FFTsize and the CIR length to the inputted CIR, thereby outputting theprocessed CIR to the second frequency domain converter (FFT2) 4121.Herein, the zero-padded CIR may correspond to one of the interpolatedCIR, extrapolated CIR, and the CIR estimated in the known data section.

The second frequency domain converter 4121 outputs the converted CIR tothe coefficient calculator 4122.

The coefficient calculator 4122 uses the frequency domain CIR beingoutputted from the second frequency domain converter 4121 to calculatethe equalization coefficient. Then, the coefficient calculator 4122outputs the calculated coefficient to the distortion compensator 4130.Herein, for example, the coefficient calculator 4122 calculates achannel equalization coefficient of the frequency domain that canprovide minimum mean square error (MMSE) from the CIR of the frequencydomain, which is outputted to the distortion compensator 4130.

The distortion compensator 4130 performs a complex number multiplicationon the overlapped data of the frequency domain being outputted from theFFT unit 4102 of the first frequency domain converter 4100 and theequalization coefficient calculated by the coefficient calculator 4122,thereby compensating the channel distortion of the overlapped data beingoutputted from the FFT unit 4102.

Block Decoder

Meanwhile, if the data being inputted to the block decoder 1303, afterbeing channel-equalized by the equalizer 1302, correspond to the datahaving both block encoding and trellis encoding performed thereon (i.e.,the data within the RS frame, the signaling information data, etc.) bythe transmitting system, trellis decoding and block decoding processesare performed on the inputted data as inverse processes of thetransmitting system. Alternatively, if the data being inputted to theblock decoder correspond to the data having only trellis encodingperformed thereon (i.e., the main service data), and not the blockencoding, only the trellis decoding process is performed on the inputteddata as the inverse process of the transmitting system.

The trellis decoded and block decoded data by the block decoder 1303 arethen outputted to the RS frame decoder 1304. More specifically, theblock decoder 1303 removes the known data, data used for trellisinitialization, and signaling information data, MPEG header, which havebeen inserted in the data group, and the RS parity data, which have beenadded by the RS encoder/non-systematic RS encoder or non-systematic RSencoder of the transmitting system. Then, the block decoder 1303 outputsthe processed data to the RS frame decoder 1304. Herein, the removal ofthe data may be performed before the block decoding process, or may beperformed during or after the block decoding process.

Meanwhile, the data trellis-decoded by the block decoder 1303 areoutputted to the data deinterleaver of the main service data processor.At this point, the data being trellis-decoded by the block decoder 1303and outputted to the data deinterleaver may not only include the mainservice data but may also include the data within the RS frame and thesignaling information. Furthermore, the RS parity data that are added bythe transmitting system after the pre-processor may also be included inthe data being outputted to the data deinterleaver.

According to another embodiment of the present invention, data that arenot processed with block decoding and only processed with trellisencoding by the transmitting system may directly bypass the blockdecoder 1303 so as to be outputted to the data deinterleaver. In thiscase, a trellis decoder should be provided before the datadeinterleaver.

More specifically, if the inputted data correspond to the data havingonly trellis encoding performed thereon and not block encoding, theblock decoder 1303 performs Viterbi (or trellis) decoding on theinputted data so as to output a hard decision value or to perform ahard-decision on a soft decision value, thereby outputting the result.

Meanwhile, if the inputted data correspond to the data having both blockencoding process and trellis encoding process performed thereon, theblock decoder 1303 outputs a soft decision value with respect to theinputted data.

In other words, if the inputted data correspond to data being processedwith block encoding by the block processor and being processed withtrellis encoding by the trellis encoding module, in the transmittingsystem, the block decoder 1303 performs a decoding process and a trellisdecoding process on the inputted data as inverse processes of thetransmitting system. At this point, the RS frame encoder of thepre-processor included in the transmitting system may be viewed as anouter (or external) encoder. And, the trellis encoder may be viewed asan inner (or internal) encoder. When decoding such concatenated codes,in order to allow the block decoder 1303 to maximize its performance ofdecoding externally encoded data, the decoder of the internal codeshould output a soft decision value.

FIG. 30 illustrates SCCC encoding process according to an embodiment ofthe present invention.

The SCCC encoding process is related with Convolutional Encoder 30010,Symbol Interleaver 30020, Symbol to Byte Converter 30030, Data MUX 30040and Trellis Encoding Module 30050.

A SCCC Decoder can decode both the main trellis code and the M/Hconvolutional code, considering that they are effectively concatenatedwith each other at the transmitter through the symbol interleaver 30020and the data mux module 30040 as shown in FIG. 30. the data mux module30040, shown as a single block, actually consists of a number ofprocessors including the Group formatter, the Packet formatter, thePacket mux, the RS encoder, the data interleaver, the byte to symbolconverter and the 12-way symbol demultiplexer in the 12-way trellisencoder.

FIG. 31 illustrates a detailed block view showing a block decoder 1303according to an embodiment of the present invention. The block decoder1303 includes an input buffer 5011, a Trellis Code Modulation (TCM)decoder 5012, a data demultiplexer 5013, a symbol deinterleaver 5014, asymbol decoder 5015, a symbol interleaver 5016, and a data multiplexer5017. The TCM decoder 5012 is referred to as an inner decoder, and thesymbol decoder 5015 is referred to as an outer decoder or a trellisdecoder. The block decoder 1303 according to the embodiment of thepresent invention performs SCCC block decoding in SCCC block units onthe inputted data. In FIG. 31, ‘U’ and ‘C’ marked on the TCM decoder5012 and the symbol decoder 5015 respectively indicate 4 ports of softinput soft output (SISO).

The input buffer 5011 temporarily stores values of mobile service datasymbols (i.e., including RS parity data symbols that were added duringRS frame encoding, and CRC data symbols) being channel-equalized andoutputted from the channel equalizer 5011 in SCCC block units.Thereafter, the input buffer 5011 repeatedly outputs the stored valuesto the TCM decoder 5012.

Also, among the symbol values being outputted from the channel equalizer1302, input symbol values of section do not include any mobile servicedata symbol (i.e., including RS parity data symbols that were addedduring RS frame encoding, and CRC data symbols) values bypass the inputbuffer 5011 without being stored. More specifically, since onlytrellis-decoding is performed on the input symbol value of sections thatare not processed with SCCC block encoding, the input buffer 5011directly outputs such input to the TCM decoder 5012 without performingany temporary storing or repeated outputting processes.

The input buffer 5011 refers to information associated to SCCC beingoutputted from the operation controller 1307 or the signaling decoder1308, e.g., the SCCC block mode and SCCC outer code mode, so as tocontrol the storage and output of the input data.

In correspondence with the 12-way trellis encoder, the TCM decoder 5012includes a 12-way Trellis Coded Modulation (TCM) decoder. Herein, 12-waytrellis-decoding is performed on the input symbol value as an inverseprocess of the 12-way trellis-encoder.

More specifically, the TCM decoder 5012 receives as many output symbolvalues of the input buffer 5011 and soft-decision values being fed-backthrough the data multiplexer 5017 as each SCCC blocks, so as to performTCM decoding on each symbol.

At this point, the soft-decision values that are fed-back are matched tobe in a one-to-one correspondence with a number of symbol positionscorresponding to the number of SCCC blocks being outputted from theinput buffer 5011, so that the matched soft-decision values can beinputted to the TCM decoder 5012 based upon the control of the datamultiplexer 5017. More specifically, the symbol values being outputtedfrom the input buffer 5011 and the turbo-decoded and inputted data arematched to one another in accordance with the same position within therespective SCCC block, thereby being outputted to the TCM decoder 5012.For example, if the turbo-decoded data correspond to the third symbolvalue within the SCCC block, the corresponding turbo-decoded data arematched with the third symbol value within the SCCC block beingoutputted from the input buffer 5011, thereby being outputted to the TCMdecoder 5012.

In order to do so, the data multiplexer 5017 controls the system so thatthe input buffer 5011 can store the corresponding SCCC block data whilethe iterative turbo decoding is being performed. And, by using a delaymethod, the data multiplexer 5017 also controls the system so that thesoft-decision value (e.g., LLR) of the output symbol of the symbolinterleaver 5016 can be matched, so as to be in a one-to-onecorrespondence, with the symbol value of the input buffer 5011corresponding to the same position (or location) within the block of theoutput symbol, thereby being inputted to the TCM decoder of thecorresponding way. At this point, in case of a symbol value that is notblock decoded, since the corresponding symbol value is not turbodecoded, a null bit is inputted in the matched output position (orlocation).

After performing this process for a predetermined number of iteration ofturbo decoding, the data of the next SCCC block is stored in the inputbuffer 5011 and then outputted, so as to repeat the turbo-decodingprocess.

The output of the TCM decoder 5012 signifies the reliability of thesymbols being inputted to the trellis encoder of the transmitting systemwith respect to the transmitted symbols. For example, since the 2-bitinput of the trellis encoding module of the transmitting systemcorresponds to one symbol, a Log Likelihood Ratio (LLR) between thelikelihood (or probability) of one bit being ‘1’ and the likelihood (orprobability) of another bit being ‘0’ may be respectively outputted(bit-unit output) for the upper bit and the lower bit. The LogLikelihood Ratio (LLR) signifies a log value on a ratio between thelikelihood value of the input bit being ‘1’ and the likelihood value ofthe input bit being ‘0’. Alternatively, a log likelihood ratio of thelikelihood value of 2 bits, i.e., one symbol being “00”, “01”, “10”, and“11” may be outputted (symbol-unit output) for all four combinations(00,01,10,11). This eventually corresponds to the soft-decision value ofthe received symbol, which indicates the reliability of the bits thatwere inputted to the trellis encoder. Herein, a Maximum A posterioriProbability (MAP), a Soft-Out Viterbi Algorithm (SOYA) may be used asthe decoding algorithm of each TCM decoder included in the TCM decoder5012.

The data demultiplexer 5013 identifies the soft-decision valuescorresponding mobile service data symbols (i.e., including RS paritydata added when performing RS frame encoding, and CRC data symbols) fromthe output of the TCM decoder 5012, thereby outputting the identifiedsoft-decision values to the symbol deinterleaver 5014. At this point,the data demultiplexer 5013 performs an inverse process of processreordering of a mobile service data symbol generated from anintermediate step, wherein the output symbols outputted from the blockprocessor of the transmitting system are being inputted to the trellisencoding module (e.g., when the symbols pass through the groupformatter, the data deinterleaver, the packet formatter, and the datainterleaver). Thereafter, the data demultiplexer 5013 performsreordering of the process order of soft decision values corresponding tothe mobile service data symbols and, then, outputs the processed mobileservice data symbols to the symbol deinterleaver 5014.

This is because a plurality of blocks exist between the block processorand the trellis encoding module, and because, due to these blocks, theorder of the mobile service data symbols being outputted from the blockprocessor and the order of the mobile service data symbols beinginputted to the trellis encoding module are not identical to oneanother. More specifically, the data demultiplexer 5013 reorders (orrearranges) the order of the mobile service data symbols being outputtedfrom the outer TCM decoder 5012, so that the order of the mobile servicedata symbols being inputted to the symbol deinterleaver 5014 matches theorder of the mobile service data symbols outputted from the blockprocessor of the transmitting system. The reordering process may beembodied as one of software, middleware, and hardware.

The symbol deinterleaver 5014 performs symbol deinterleaving on the softdecision values of data symbols being reordered and outputted from thedata demultiplexer 5013 as an inverse process of the symbol interleaverincluded in the transmitting system. The size of the SCCC block beingused by the symbol deinterleaver 5014, during the symbol deinterleavingprocess, is identical to the interleaving size (i.e., B) of an actualsymbol of the symbol interleaver included in the transmitting system.This is because turbo decoding is performed between the TCM decoder 5012and the symbol decoder 5015.

The input and output of the symbol interleaver 5014 all corresponds tosoft-decision values, and the deinterleaved soft-decision values areoutputted to the symbol decoder 5015.

The symbol decoder 5015 has 4 memory states. If the symbol decoder is ina ½ coding rate mode, the memory states are changed in accordance withan input LLR set respective to a symbol. More specifically, in case ofdata being ½-rate encoded and outputted, the output LLR of the symboldeinterleaver 5014 is directly outputted to the symbol decoder 5015.

However, if the symbol decoder is in a ¼ coding rate mode, i.e., in caseof data being ¼-rate encoded and outputted from the symbol encoder ofthe transmitting system, the memory states are changed in accordancewith 2 input LLR sets respective to 2 successive symbols. Therefore, 2input LLR sets should be merged into one LLR set during the input stageof the symbol decoder 5015. In the present invention, the merged LLR setmay be obtained by adding first input LLRs and second input LLRs. IfLi(x) is defined as an input LLR value having a condition of ‘x’, themerged LLR set may be expressed by using Equation 8 shown below.

Li(merged nibble=‘0000’)=Li(first symbol=‘00’)+Li(second symbol=‘00”)

Li(merged nibble=‘0001’)=Li(first symbol=‘00’)+Li(second symbol=‘01”)

Li(merged nibble=‘0010’)=Li(first symbol=‘00’)+Li(second symbol=‘10”)

Li(merged nibble=‘0011’)=Li(first symbol=‘00’)+Li(second symbol=‘11”)

Li(merged nibble=‘0100’)=Li(first symbol=‘01’)+Li(second symbol=‘00”)

Li(merged LLR=‘1111’)=Li(first symbol=‘11’)+Li(secondsymbol=‘11”)  Equation 8

Meanwhile, as the opposite of the input LLR processing, the processingof the LLR that is to be outputted from the symbol decoder 5015 isdivided into 2 symbol LLRs in the ¼-code rate mode, as shown in Equation9 below, thereby being outputted.

Lo(first symbol=‘00’)=Maximum Probability whose LLR is from the sets{Lo(merged nibble=‘00XY’)+Delta}

Lo(first symbol=‘01’)=Maximum Probability whose LLR is from the sets{Lo(merged nibble=‘01XY’)+Delta}

Lo(first symbol=‘10’)=Maximum Probability whose LLR is from the sets{Lo(merged nibble=‘10XY’)+Delta}

Lo(first symbol=‘11’)=Maximum Probability whose LLR is from the sets{Lo(merged nibble=‘00XY’)+Delta}

Lo(second symbol=‘00’)=Maximum Probability whose LLR is from the sets{Lo(merged nibble=‘XY00’)+Delta}

Lo(second symbol=‘01’)=Maximum Probability whose LLR is from the sets{Lo(merged nibble=‘XY01’)+Delta}

Lo(second symbol=‘10’)=Maximum Probability whose LLR is from the sets{Lo(merged nibble=‘XY10’)+Delta}

Lo(second symbol=‘1’)=Maximum Probability whose LLR is from the sets{Lo(merged nibble=‘XY00’)+Delta}  Equation 9

Herein, X and Y are the arbitrary selected digits from digit 0 or 1.Also, according to an embodiment of the present invention, a correctionterm ‘Delta’ value is calculated from an IETF RFC 3926 “FLUTE—FileDelivery over Unidirectional Transport”.

At this point, the symbol decoder 5015 output 2 types of soft-decisonvalues. One corresponds to a soft-decision value being matched with anoutput symbol of the symbol decoder (hereinafter referred to as a firstsoft-decision value). And, the other corresponds to a soft-decisionvalue being matched with an input symbol of the symbol decoder(hereinafter referred to as a second soft-decision value). The firstsoft-decision value represents a reliability of the output symbol, i.e.,two bits, of the symbol encoder. And, a Log Likelihood Ratio (LLR)between the likelihood (or probability) of one bit being ‘1’ and thelikelihood (or probability) of another bit being ‘0’ may be respectivelyoutputted (bit-unit output) for the upper bit and the lower bit, whichconfigure a symbol. Alternatively, a log likelihood ratio of thelikelihood value of 2 bits, i.e., one symbol being “00”, “01”, “10”, and“11” may be outputted (symbol-unit output) for all combinations. Thefirst soft-decision value is fed-back to the TCM decoder 5012 throughthe symbol interleaver 5016 and the data multiplexer 5017. The secondsoft-decision value represents a reliability of the input symbol of thesymbol encoder of the transmitting system. Herein, the secondsoft-decision value is expressed as a Log Likelihood Ratio (LLR) betweenthe likelihood (or probability) of one bit being ‘1’ and the likelihood(or probability) of another bit being ‘0’, thereby being outputted tothe RS frame decoder 1304. Herein, a Maximum A posteriori Probability(MAP), a Soft-Out Viterbi Algorithm (SOYA) may be used as the decodingalgorithm of the symbol decoder 5015.

At this point, when the first soft-decision value being outputted fromthe symbol decoder 5015 is in a ¼ coding rate mode, the firstsoft-decision value is divided into 2 symbol LLRs, as shown in Equation9, so as to be outputted to the symbol interleaver 5016.

For example, when the input/output unit of the symbol decoder 5015corresponds to symbol units, 16 (2⁴=16) different types of soft-decisionvalues (LLRs) are inputted to the symbol decoder 5015. At this point,the 16 (2⁴=16) different types of soft-decision values (i.e., LLRs)being inputted to the symbol decoder 5015 correspond to results ofadding the respective first input LLR and the respective second inputLLR.

If ¼-rate coding is performed by the symbol encoder, the symbol decoder5015 receives the LLR respective to the 16 different symbols andperforms symbol decoding. Thereafter, the symbol decoder 5015 may outputthe LLR respective to the 16 different symbols as the firstsoft-decision value. Alternatively, the symbol decoder 5015 may receivethe LLR respective to 4 bits and performs symbol decoding. Thereafter,the symbol decoder 5015 may output the LLR respective to the 4 bits asthe first soft-decision value.

And, if ½-rate coding is performed by the symbol encoder, the symboldecoder 5015 receives the LLR respective to the 4 different symbols andperforms symbol decoding. Thereafter, the symbol decoder 5015 may outputthe LLR respective to the 4 different symbols as the first soft-decisionvalue. Alternatively, the symbol decoder 5015 may receive the LLRrespective to 2 bits and performs symbol decoding. Thereafter, thesymbol decoder 5015 may output the LLR respective to the 2 bits as thefirst soft-decision value.

According to an embodiment of the present invention, the symbolinterleaver 5016 performs symbol interleaving on the first soft-decisionvalue being outputted from the symbol decoder 5015, as shown in FIG. 32,thereby outputting the symbol-interleaved first soft-decision value tothe data multiplexer 5017. Herein, the output of the symbol interleaver5020 also becomes a soft-decision value. According to another embodimentof the present invention, the symbol interleaver 5016 performs symbolinterleaving on the first soft-decision value being outputted from thesymbol decoder 5015, as shown in FIG. 32, thereby outputting thesymbol-interleaved first soft-decision value to the data multiplexer5017.

If the SCCC block mode is ‘00’, a data group is configured of 10 SCCCblocks. And, if the SCCC block mode is ‘01’, a data group is configuredof 5 SCCC blocks. At this point, the symbol interleaving pattern of the15 SCCC blocks are different from one another. Therefore, in order tostore all symbol interleaving patterns, a memory having a very largecapacity (e.g., ROM) is required. FIG. 32 illustrates a block viewshowing the structure of a symbol interleaver according to the presentinvention, wherein the symbol interleaver can perform symbolinterleaving without requiring a memory, such as ROM. More specifically,when the SCCC block is inputted, symbol interleaved data may be directlyoutputted without having to use a memory, such as ROM.

The symbol interleaver 5016 of FIG. 32 includes a pattern generator 5110and a pattern output unit 5220. The pattern generator 5110 may include amodulo counter 5111, a multiplexer 5113, an accumulator 5114, amultiplier 5115, and a modulo operator 5116. The pattern output unit5220 may include a data remover 5221 and a buffer 5222. Herein, a modulooperation may be further included between the accumulator 5114 and themultiplier 5115. Also, the multiplier 5115 may be configured of multipleadders (or shifters).

In FIG. 32, B represents a Block length in symbols (e.g., SCCC blocklength) being inputted for symbol interleaving. And, L corresponds to asymbol unit block length actually being interleaved y the symbolinterleaver 5016. At this point, L=2m (wherein m is an integer), whereinL should satisfy the condition of L>B.

The modulo counter 5111 performs sequential counting starting from 0 toL. The accumulator 5114 receives a count value of the modulo counter5111 starting from 0. The multiplexer 5113 selects a constant whenstarting the symbol interleaving process on an SCCC block. Thereafter,the multiplexer 5113 is fed-back with the output of the accumulator5114, thereby outputting the feedback to the accumulator 5114. In thiscase, an initial offset value of symbol interleaving is equal to 0.

The accumulator 5114 adds the output of the modulo counter 5111 and theoutput of the multiplexer 5113 and, then outputs the added value to themultiplier 5113.

The multiplier 5115 multiplies the output of the accumulator 5114 by aconstant 89, thereby outputting the multiplied result to the modulooperator 5116. The modulo operator 5116 performs a modulo L operation onthe output of the multiplier 5115, thereby outputting the processed datato the pattern output unit 5220. According to an embodiment of thepresent invention, the modulo operator 5116 uses a bitwise mask functionto perform the modulo L operation. For example, when the L value isequal to 210, and when only the lower 10 bits among the output of themultiplier 5115 are outputted from the modulo operator 5116 and inputtedto the pattern output unit 5220, the modulo L operation is performed.

When the output value is equal to or greater than B, the data remover5221 of the pattern output unit 5220 discards the input value andoutputs the processed data to the buffer 5222. According to anembodiment of the present invention, the buffer 5222 is configured tohave a First Input First Output (FIFO) structure. The buffer 5222condenses the remaining values that have not been discarded by the dataremover 5221 and then stores the condensed values, which are thenoutputted in accordance with the stored order. Therefore, the firstoutput B outputted from the buffer 5222 corresponds to the symbolinterleaving pattern P(i). At this point, the index i value of thesymbol interleaving pattern P(i) increases from 0 to B−1. If the modulocounter 5111 continues to be operated, and when the next output B iscollected from the buffer 5222, the symbol interleaving pattern at thispoint becomes the inverse order of the symbol interleaving pattern P(i).More specifically, the index i value of the symbol interleaving patternP(i) decreases from B−1 to 0.

Therefore, when the initial offset is set to an L/2 value and not to‘0’, and when symbol interleaving is started, the first B output becomesan inverse order of the interleaving pattern P(i). In this case, theinitial offset value of symbol interleaving becomes an L/2 value.

If 0 is used as the initial offset value, the Lth value being fed-backfrom the accumulator 5114 becomes (L−1)*L/2, and the modulo L of thefeedback value is L/2.

For example, when the initial offset value is set to 0, the symbolinterleaving pattern P(i) may be obtained. And, when the initial offsetvalue is set to an L/2 value, an inverse order of the interleavingpattern P(i), i.e., a symbol deinterleaving pattern P(i)−1 may beobtained from the beginning. For example, when the symbol deinterleaver5014 sets an L/2 value as the initial offset value, and when the symbolinterleaver 5016 sets ‘0’ as the initial offset value, only one symbolinterleaving pattern P(i) is used to performed both the symboldeinterleving and symbol interleaving processes.

Alternatively, when only one initial offset is set, and when the modulooperator 5111 performs a counting process up to 2L, a symbolinterleaving pattern and a symbol deinterleaving pattern may begenerated by using a single initial offset.

FIG. 33 illustrates an example of a symbol interleaving patterngenerated when the offset value is equal to 0 according to the presentinvention. In the example shown in FIG. 33, L is equal to 12000, and theSCCC block length is equal to 16384. Herein, the output pattern in anindex starting from 12000 to 23999 corresponds to an inverse order ofthe output pattern in an index starting from 0 to 11999.

Also, since interleaving and deinterleaving are inverse processes of oneanother, the interleaving pattern P(i) and the deinterleaving patternP(i)−1 are not required to be separately generated by the block decoder1303. More specifically, symbol interleaving and deinterleavingoperations may both be performed by using only the symbol interleavingpattern P(i).

(a) of FIG. 34 shows an exemplary process of performing symbolinterleaving by using only the symbol interleaving pattern P(i). And,(b) of FIG. 34 shows an exemplary process of performing symboldeinterleaving by using only the symbol interleaving pattern P(i).

In (a) of FIG. 34, the symbol interleaving process is as describedbelow.

1a. An interleaving pattern P(i) is generated.

1b. The ith input data symbol is written in location i of the memory.

1c. Starting from location i of the memory, an ith output data symbol isread.

When the processes 1a to 1c are performed from 0 to B−1, the symbolinterleaving process for one SCCC block is completed. Herein, the memorymay correspond to a buffer 5222.

In (b) of FIG. 34, the symbol deinterleaving process is as describedbelow.

2a. An interleaving pattern P(i) is generated.

2b. The ith input data symbol is written in location i of the memory.

2c. Starting from location P(i) of the memory, an ith output data symbolis read.

When the processes 2a to 2c are performed from 0 to B−1, the symboldeinterleaving process for one SCCC block is completed. Herein, thevalue of i ranges from 0 to B−1.

More specifically, in (a) and (b) of FIG. 34, step 1b and step 2c accessthe same address of the memory, and step 1c and step 2b access the sameaddress of the memory.

Therefore, after reading previous data starting from a specific location(or position) of the memory, when current data are written in thecorresponding location (or position), the symbol interleaver 5016 andthe symbol deinterleaver 5014 may use a single permutation memory so asto perform symbol interleaving and symbol deinterleaving. Morespecifically, since an address of the memory can be shared during thesymbol interleaving and symbol deinterleaving processes, the memory sizemay be reduced.

As described above, in the present invention, only one symbolinterleaving pattern is used to perform symbol interleaving and symboldeinterleaving, thereby having the effect of reducing the memory size.

More specifically, the data multiplexer 5017 of the block decoder 1303reorders (or rearranges) the output order of the symbol interleaver 5016in accordance with the processing order of the symbol generated from anintermediate step (e.g., the group data formatter, the packet formatter,the data interleaver). Thereafter, the data multiplexer 5017 outputs theprocessed symbols to the TCM decoder 5012. Herein, the reorderingprocess of the data multiplexer 5017 may be embodied as at least one ofsoftware, middleware, and hardware.

The soft-decision values being outputted from the symbol interleaver5016 are matched to be in a one-to-one correspondence with mobileservice data symbol positions corresponding to the number of SCCC blocksbeing outputted from the input buffer 5011. Then, the matchedsoft-decision values are inputted to the TCM decoder 5012. At thispoint, since a main service data symbol or an RS parity symbol, knowndata symbol, signaling information data, and so on, of the main servicedata do not correspond to mobile service data symbols, the datamultiplexer 5017 inserts null data in the corresponding location (orposition), thereby outputting the processed data to the TCM decoder5012. Also, each time the symbols of the SCCC blocks are turbo-decoded,since there is no value being fed-back from the symbol interleaver 5016at the beginning of the first decoding process, the data multiplexer5017 inserts null data in all symbol positions including a mobileservice data symbol, thereby transmitting the processed data to the TCMdecoder 5012.

The second soft-decision values being outputted from the symbol decoder5015 are inputted to the RS frame decoder 1304. For example, the symboldecoder 5015 does not output any second soft-decision value until turbodecoding is performed for a predetermined number of repetition (oriteration) times (e.g., M number of times). Thereafter, when M number ofturbo-decoding processes on one SCCC block is all performed, the secondsoft-decision value of that specific point is outputted to the RS Framedecoder 1304. More specifically, after performing turbo-decoding for apredetermined number of times, the soft decision value of the symboldecoder 5015 is outputted to the RS frame decoder 1304. And, thus, theblock decoding process on one SCCC block is completed.

In the present invention, this will be referred to as an iterative turbodecoding process for simplicity.

At this point, the number of iterative turbo decoding performed betweenthe TCM decoder 5012 and the symbol decoder 5015 may be defined byconsidering hardware complexity and error correction performance.Accordingly, when the number of iterative turbo decoding increases, theerror correction can be enhanced. However, this case disadvantageous inthat the hardware may also increase.

RS Frame Decoder

FIG. 35 illustrates a structure of an RS frame decoder according to anembodiment of the present invention.

RS frame decoder includes RS Frame builder 6111, RS-CRC Decoder 6112 andM/H TP Interface block 1305.

As shown in FIG. 35, an RS Frame decoder processes a particular M/HensembleEnsemble selected by upper layer request. An RS Frame Buildercollects data from the selected Ensemble and builds an RS framecorresponding to the selected Ensemble. An RS-CRC decoder detects andcorrects errors in the completed RS frame. An M/H TP interface blockderandomizes the data to undo the effects of the M/H randomizer at thetransmitter, and finally outputs M/H TPs.

A RS Frame Decoder builds an RS Frame, detects errors on each row of theRS frame by CRC decoding, corrects errors by erasure RS decoding witherror location information from CRC decoding and SCCC decoding on eachcolumn of the RS frame, and outputs M/H TPs (Transport Packets) withmarked error indication fields.

FIG. 36 illustrates, when the RS frame mode value is equal to ‘00’, anexemplary process of grouping several portion being transmitted to aparade, thereby forming an RS frame and an RS frame reliability map.

More specifically, the RS frame decoder 2006 receives and groups aplurality of mobile service data bytes, so as to form an RS frame.According to the present invention, in transmitting system, the mobileservice data correspond to data RS-encoded in RS frame units. At thispoint, the mobile service data may already be error correction encoded(e.g., CRC-encoded). Alternatively, the error correction encodingprocess may be omitted.

It is assumed that, in the transmitting system, an RS frame having thesize of (N+2)×(187+P) bytes is divided into M number of portions, andthat the M number of mobile service data portions are assigned andtransmitted to regions A/B/C/D in M number of data groups, respectively.In this case, in the receiving system, each mobile service data portionis grouped, as shown in FIG. 36( a), thereby forming an RS frame havingthe size of (N+2)×(187+P) bytes. At this point, when stuffing bytes (S)are added to at least one portion included in the corresponding RS frameand then transmitted, the stuffing bytes are removed, therebyconfiguring an RS frame and an RS frame reliability map. For example,when S number of stuffing bytes are added to the corresponding portion,the S number of stuffing bytes are removed, thereby configuring the RSframe and the RS frame reliability map.

Herein, when it is assumed that the block decoder 1303 outputs a softdecision value for the decoding result, the RS frame decoder 1304 maydecide the ‘0’ and ‘1’ of the corresponding bit by using the codes ofthe soft decision value. 8 bits that are each decided as described aboveare grouped to create 1 data byte. If the above-described process isperformed on all soft decision values of several portions (or datagroups) included in a parade, the RS frame having the size of(N+2)×(187+P) bytes may be configured.

Additionally, the present invention uses the soft decision value notonly to configure the RS frame but also to configure a reliability map.

Herein, the reliability map indicates the reliability of thecorresponding data byte, which is configured by grouping 8 bits, the 8bits being decided by the codes of the soft decision value.

For example, when the absolute value of the soft decision value exceedsa pre-determined threshold value, the value of the corresponding bit,which is decided by the code of the corresponding soft decision value,is determined to be reliable. Conversely, when the absolute value of thesoft decision value does not exceed the pre-determined threshold value,the value of the corresponding bit is determined to be unreliable.Thereafter, if even a single bit among the 8 bits, which are decided bythe codes of the soft decision value and group to configure one databyte, is determined to be unreliable, the corresponding data byte ismarked on the reliability map as an unreliable data byte.

Herein, determining the reliability of one data byte is only exemplary.More specifically, when a plurality of data bytes (e.g., at least 4 databytes) are determined to be unreliable, the corresponding data bytes mayalso be marked as unreliable data bytes within the reliability map.Conversely, when all of the data bits within the one data byte aredetermined to be reliable (i.e., when the absolute value of the softdecision values of all 8 bits included in the one data byte exceed thepredetermined threshold value), the corresponding data byte is marked tobe a reliable data byte on the reliability map. Similarly, when aplurality of data bytes (e.g., at least 4 data bytes) are determined tobe reliable, the corresponding data bytes may also be marked as reliabledata bytes within the reliability map. The numbers proposed in theabove-described example are merely exemplary and, therefore, do notlimit the scope or spirit of the present invention.

The process of configuring the RS frame and the process of configuringthe reliability map both using the soft decision value may be performedat the same time. Herein, the reliability information within thereliability map is in a one-to-one correspondence with each byte withinthe RS frame. For example, if a RS frame has the size of (N+2)×(187+P)bytes, the reliability map is also configured to have the size of(N+2)×(187+P) bytes. FIG. 36( a′) and FIG. 36( b′) respectivelyillustrate the process steps of configuring the reliability mapaccording to the present invention.

Subsequently, the RS frame reliability map is used on the RS frames soas to perform error correction.

FIG. 37 and FIG. 38 illustrate an error correction decoding processaccording to an embodiment of the present invention.

According to an embodiment of the present invention, in case of FIG. 37,a CRC syndrome check process is performed once again on the CRC-RSdecoded RS frame. And, the result of the CRC syndrome check process ismarked in an error_indicator field within each M/H service data packetconfiguring the payload of the RS frame. Thereafter, the marked resultis outputted for A/V decoding. For example, the error_indicator field ofthe M/H service data packet having an error existing therein is markedas ‘1’, and the error_indicator field of the M/H service data packethaving no error existing therein is marked as ‘0’. According to theembodiment of the present invention, if the error_indicator field valueof all M/H service data packets within the RS frame payload is set to‘0’ and transmitted by the transmitting system, then based upon the CRCsyndrome check result, only the error_indicator fields of the M/Hservice data packet rows are marked as ‘1’.

Thus, the probability of malfunctioning in blocks receiving andprocessing M/H service data packets (e.g., M/H TP interface block 1305)in later processes may be reduced. For example, the M/H TP interfaceblock 1305 may discard any M/H service data packet having theerror_indicator field marked as ‘1’ without using the corresponding M/Hservice data packet. Accordingly, since the probability ofmalfunctioning in the M/H TP interface block 1305 can be reduced, theoverall performance of the receiving system may be enhanced.

More specifically, when a (N+2)×(187+P)-byte size RS frame and a(N+2)×(187+P)-bit size RF frame reliability map are configured, as shownin (a) and (a′) of FIG. 37, a CRC syndrome check is performed on the RSframe, so as to check whether or not an error has occurred in each row.Subsequently, the presence or absence of an error is marked on a CRCerror flag corresponding to each row, as shown in (b) of FIG. 37. Atthis point, since the portion of the reliability map corresponding tothe CRC checksum as no applicability, the corresponding portion isremoved (or deleted or discarded), so that only N×(187+P) number ofreliability information remains, as shown in (b′) of FIG. 37.

As described above, after performing the CRC syndrome check, (187+P,187)-RS decoding is performed on N number of columns At this point,RS-decoding is performed on only N number of columns excluding the last2 columns from the overall (N+2) number of columns because the last 2columns are configured only of CRC checksum and also because thetransmitting system did not perform RS-encoding on the last 2 columns

At this point, depending upon the number of errors marked on the CRCerror flag, either an erasure decoding process is performed or a generalRS-decoding process is performed.

For example, when the number of rows including CRC error is less than orequal to a maximum number of errors correctable by RS erasure decoding(according to the embodiment of the present invention, the maximumnumber is ‘48’), (235,187)-RS erasure decoding is performed on the RSframe having (18+P) number of N-byte rows, i.e., the RS frame having 235N-byte rows in a column direction, as shown in (d) of FIG. 37. However,when the number of rows including CRC error is greater than the maximumnumber of errors (i.e., 48 errors) correctable by RS erasure decoding,RS erasure decoding cannot be performed. In this case, error correctionmay be performed through a general RS-decoding process. Herein, thepresent invention may further enhance the error correcting ability byusing the reliability map, which was generated when configuring the RSframe, from a soft decision value.

More specifically, the RS frame decoder compares the absolute value ofthe soft decision value of the block decoder 1303 with thepre-determined threshold value, so as to determine the reliability ofthe bit value decided by the code of the corresponding soft decisionvalue. Also, 8 bits, each being determined by the code of the softdecision value, are grouped to form one data byte.

Accordingly, the reliability information on this one data byte isindicated on the reliability map. Therefore, as shown in FIG. 37( c),even though a particular row is determined to have an error occurringtherein based upon a CRC syndrome checking process on the particularrow, the present invention does not assume that all bytes included inthe row have errors occurring therein. The present invention refers tothe reliability information of the reliability map and sets only thebytes that have been determined to be unreliable as erroneous bytes. Inother words, with disregard to whether or not a CRC error exists withinthe corresponding row, only the bytes that are determined to beunreliable based upon the reliability map are set as erasure points.

According to another method, when it is determined that CRC errors areincluded in the corresponding row, based upon the result of the CRCsyndrome checking result, only the bytes that are determined by thereliability map to be unreliable are set as errors. More specifically,only the bytes corresponding to the row that is determined to haveerrors included therein and being determined to be unreliable based uponthe reliability information, are set as the erasure points.

Thereafter, if the number of error points for each column is smallerthan or equal to the maximum number of errors (i.e., 48 errors) that canbe corrected by the RS erasure decoding process, an RS erasure decodingprocess is performed on the corresponding column Conversely, if thenumber of error points for each column is greater than the maximumnumber of errors (i.e., 48 errors) that can be corrected by the RSerasure decoding process, a general decoding process is performed on thecorresponding column

More specifically, if the number of rows having CRC errors includedtherein is greater than the maximum number of errors (i.e., 48 errors)that can be corrected by the RS erasure decoding process, either an RSerasure decoding process or a general RS decoding process is performedon a column that is decided based upon the reliability information ofthe reliability map, in accordance with the number of erasure pointswithin the corresponding column

For example, it is assumed that the number of rows having CRC errorsincluded therein within the RS frame is greater than 48. And, it is alsoassumed that the number of erasure points decided based upon thereliability information of the reliability map is indicated as 40erasure points in the first column and as 50 erasure points in thesecond column. In this case, a (235,187)-RS erasure decoding process isperformed on the first column. Alternatively, a (235,187)-RS decodingprocess is performed on the second column

As described above, the present invention may apply the process (d) ofFIG. 37 or the process (d′) of FIG. 37, so as to perform errorcorrection decoding on N number of columns excluding the last 2 columnswithin the RS frame.

After performing error correction decoding on the N number of columns,the number of RS errors is counted as shown in (e) of FIG. 38

At this point, if an error did not occur in any of the columns, or ifall errors have been corrected in process (d) of FIG. 37 or process (d′)of FIG. 37, i.e., if the number of RS errors is equal to ‘0’, thisindicates that there is no error in the (N+187)-byte RS frame payloadconfiguring the M/H service data packet within the corresponding RSframe. Herein, as shown in (f) of FIG. 38, derandomizing is performed onthe (N+187)-byte RS frame payload as an inverse process of thetransmitting system. Thereafter, when outputting each M/H service datapacket (i.e., M/H TP packet) of the derandomized RS frame payload to theM/H TP interface block 1305, the output is performed by setting thevalue of the error_indicator field within the M/H service data packet to‘0’ (i.e., indicating that there is no error), as shown in (g) of FIG.38. More specifically, the value of the error_indicator field withineach of the M/H service data packets configuring the RS frame payload isequally set to ‘0’.

Meanwhile, even though RS-decoding is performed, errors in N number ofcolumns may all remain without being corrected. In this case, the numberof RS errors is not equal to ‘0’.

In this case, as shown in (h) of FIG. 37, a CRC syndrome check isperformed once again on the RS-decoded RS frame, thereby checking onceagain whether or not an error exists in 187 rows.

The CRC syndrome check is repeated in (h) of FIG. 38 because, althoughRS-decoding has not been performed on the last 2 columns (i.e., CRCchecksum data) of the RS frame, RS-decoding has been performed on the Nnumber of columns including M/H service data packet. Accordingly, theeffects (or influence) of the error corrected by RS-decoding may beverified and reflected (or applied).

More specifically, after performing CRC-RS decoding, when the presentinvention repeats the CRC syndrome check process once again on each row,as shown in (h) of FIG. 38, and derandomizes the RS frame payloadprocessed with CRC syndrome checking, as shown in (i) of FIG. 38, andwhen the present invention outputs the derandomized RS frame payload,the present invention marks the CRC syndrome check result in theerror_indicator field of the M/H service data packet configuring thecorresponding row, as shown in (j) of FIG. 38.

For example, when performing the CRC syndrome check once again, if it isdetermined that there is not CRC error in the RS frame, the value of theerror_indicator field within each M/H service data packet of thederandomized RS frame payload is equally set to ‘0’.

When performing the CRC syndrome check once again, if it is determinedthat a CRC error exists in a specific row of the RS frame, for example,the second and third rows of the RS frame, the values of theerror_indicator field within the second and third M/H service datapackets of the derandomized RS frame payload are marked to be equal to‘1’, and the value of the error_indicator field within the remaining M/Hservice data packets is equally marked to be equal to ‘0’.

The present invention is provided with a number (=M) of RS framedecoders aligned in parallel, wherein the number corresponds to thenumber of parades included in one M/H frame. Herein, the RS framedecoder may be configured by being provided with a multiplexer connectedto the input end of each of the M number of RS frame decoders, so as tomultiplex a plurality of portions, and a demultiplexer connected to theoutput end of each of the M number of RS frame decoders.

Signaling Decoding

The signaling decoder 1306 extracts and decodes signaling information(e.g., TPC and FIC information), which was inserted and transmitted bythe transmitting system, from the received (or inputted) data.Thereafter, the signaling decoder 1306 provides the decoded signalinginformation to the block(s) requiring such information.

More specifically, the signaling decoder 1306 extracts and decodes TPCdata and FIC data, which were inserted and transmitted by thetransmitting system, from the equalized data. Then, the signalingdecoder 1306 outputs the TPC data to the operation controller 1307, andthe signaling decoder 1306 outputs the FIC data to the FIC processor1308. For example, the TPC data and the FIC data are inserted in thesignaling information region of each data group, thereby being received.

At this point, the signaling information area within the data group maybe known by using the known data position information that is outputtedfrom the known sequence detector 1506. The signaling information areacorresponds to the area starting from the first segment to a portion ofthe second segment of M/H block B4 within the data group, as shown inFIG. 4. More specifically, in the present invention, 276 (=207+69) bytesof the M/H block B4 within each data group are allocated to an area forinserting the signaling information. In other words, the signalinginformation area is configured of 207 bytes corresponding to the firstsegment of M/H block B4 and of the first 69 bytes of the second segmentof M/H block B4. Additionally, the first known data sequence (i.e.,first training sequence) is inserted in the last 2 segments of M/H blockB3, and the second known data sequence (i.e., second training sequence)is inserted in the second and third segments of M/H block B4. At thispoint, since the second known data sequence is inserted after thesignaling information area and then received, the signaling decoder 1306may extract and decode signaling information of the signalinginformation area from the data being outputted from the channelsynchronizer 1301 or the channel equalizer 1302.

FIG. 39 illustrates a block view of the signaling decoder 1306 accordingto an embodiment of the present invention. The signaling decoder 1306performs iterative turbo decoding and RS-decoding on the data of thesignaling information region among the equalized data. Thereafter, thetransmission parameter (i.e., TPC data) obtained as a result of theabove-described process is outputted to the operation controller 1307,and the FIC data are outputted to an upper layer.

For this operation, the signaling decoder 1036 may include an iterativeturbo decoder 7111, a derandomizer 7112, a demultiplexer 7113, an RSdecoder 7114, a block deinterleaver 7115, and an RS decoder 7116.

FIG. 40 is a detailed block diagram illustrating the iterative turbodecoder 7111. Referring to FIG. 40, upon receiving the signalinginformation area's data from among the equalized data, a demultiplexer(DeMux) 7200 discriminates symbols corresponding to respective branchesof the signaling encoder of the transmission system, and outputs thediscriminated symbols to buffers 7201 and 7401, respectively.

The buffers 7201 and 7401 store input data corresponding to thesignaling information area, and respectively repeatedly output thestored input data to the demultiplexers 7202 and 7402 during the turbodecoding process.

In accordance with one embodiment of the present invention, it isassumed that output data of the even encoder in the signaling encoder ofthe transmission system is processed to be input to 0^(th), 2^(nd), . .. , 10^(th) trellis encoders (i.e., even number trellis encoders), andoutput data of the odd encoder 575 is processed to be input to 1^(st),11^(th) trellis encoders. In this case, the demultiplexer 7202 outputsoutput data of the buffer 7201 to a trellis decoder (i.e., TCM decoder)corresponding to the even number trellis encoder. The demultiplexer 7202receives data fed back from the block deinterleaver 7507, and outputsthe feed-back data to the same trellis decoder (i.e., TCM decoder)corresponding to the even number trellis encoder.

In this case, output data of each trellis decoder (TCM decoder)corresponds to a log likelihood ratio (LLR) value. The LLR value is aresult from taking a logarithm of a soft decision value. Morespecifically, the LLR value corresponds to a log likelihood ratio (LLR)between a likelihood of input bit being equal to ‘1’ and a likelihood ofinput bit being equal to ‘0’. An initial value of the LLR is set tozero. The LLR value is transferred to the even component decodercorresponding to the even component encoder contained in the signalingencoder of the transmission system. Input/output (I/O) data of the evencomponent decoder is such an LLR value as well. In this case, since asingle even number trellis decoder interoperates with a single evencomponent decoder, an even component encoder and an even number trellisencoder are considered as a single encoder (effective componentencoder). Hence, the even number trellis decoder and the even componentdecoder can be merged into a single effective component decoder. In thecase where the two decoders configure a single decoder, decodingperformance will be enhanced although complexity increases due to theincreased number of states.

Output signals of the even component decoders 7300 to 7305 aresequentially transferred to the multiplexer 7306 and are thentransferred to the block interleaver 7307. The block interleaver 7307has the same configuration as a block interleaver used for the signalingencoder of the transmitting side.

The LLR value block-interleaved by the block interleaver 7307 is fedback to the demultiplexer 7402. The demultiplexer 7402 outputs the LLRvalue to a corresponding trellis decoder (i.e., TCM decoder) from amongsix trellis decoders, and at the same time transmits output data of thebuffer 2401 to the trellis decoder. For example, provided that the LLRvalue fed back from the block deinterleaver 7507 is an LLR value of thefirst decoder 7500, the demultiplexer 7402 outputs this feed-back LLRvalue and the output data of the buffer 7401 to the trellis decoder ofthe first decoder 7500.

The above-mentioned rules are equally applied to the demultiplexer 7202.The odd number trellis decoder and the odd component decoder can beoperated in the same manner as in the even number trellis decoder andthe even component decoder. Likewise, the odd number trellis decoder andthe odd component decoder can be implemented as a single effectivecomponent decoder.

Output signals of the odd number decoders 7500 to 7505 are sequentiallytransferred to the multiplexer 7506, and are then forwarded to the blockdeinterleaver 7507. The block deinterleaver 7507 is an inverse processof the block interleaver. Thus, the LLR value block-deinterleaved by theblock deinterleaver 7507 is input to the demultiplexer 7202 toaccomplish the iterative turbo decoding.

After the iterative turbo decoding has been repeatedly performed at apredetermined level, the iterative turbo-decoded result is output to thederandomizer 7112.

At this point, in the above-mentioned iterative turbo decoding process,the even and odd decoders must have trellis diagram information of acorresponding encoder. Each of the encoders shown in FIGS. 41( a) and41(b) has five memories D0 to D4 so as to obtain 32 states (i.e., 2⁵states). However, the number of states acquired when start states of allthe signaling information areas are constant may be limited to thenumber of only some states among a total of 32 states. That is, if it isassumed that a start state of the effective component encoder is limitedto a specific state, the effective component encoder may have a smallernumber of states as compared to 32 states.

For example, all memories of the even/odd component encoders of theiterative turbo encoder (i.e., PCCC encoder) are each set to zero at thebeginning of each signaling information area of a single data group.Because the signaling information area just follows a first known datasequence (i.e., 1^(st) training sequence) and the first known datasequence is designed to allow all memories in each of the twelve trellisencoders to have a state of zero at the end of the first known datasequence. As a result, the respective memories of the effectivecomponent encoder always start from a state ‘00000’. That is, allmemories of the effective component encoder are each set to a state ofzero at the beginning of the signaling information area. In this way,provided that all memories of the effective component encoders in thesignaling information area start from the state ‘00000’, the dataencoding can be achieved using only specific states among 32 statesalthough data of the signaling information area is considered to berandom.

The signaling information area ranges from a first segment of an M/Hblock 134′ of a data group to some parts of a second segment thereof.That is, 276 (=207+69) bytes of the M/H block 134′ of each data groupare assigned to an area for inserting signaling information. In otherwords, the signaling information area is composed of 207 bytescorresponding to a first segment of the M/H block 134′ and first 69bytes of a second segment thereof. In addition, the first known datasequence (i.e., the first training sequence) is inserted into the last 2segments of an M/H block 133′, and a second known data sequence (i.e.,the second training sequence) is inserted into second and third segmentsof an M/H block 134′. In this case, the second known data sequence islocated just behind the signaling information area. Third to sixth knowndata sequences (i.e., third to sixth training sequences) arerespectively inserted into the last 2 segments of the M/H blocks B4, B5,B6, and B7.

FIG. 41( a) illustrates an exemplary case in which a trellis encoder isserially concatenated with the even component encoder.

In fact, although a plurality of blocks are located between the evencomponent encoder and the trellis encoder, the receiving systemconsiders two blocks to be concatenated with each other, so that itdecodes data. In other words, the trellis encoder performs precoding onthe high-order bit ‘X2’ generated from the even component encoder, andoutputs the precoded result as a most significant bit ‘Z2’. In addition,the trellis encoder performs trellis-encoding on the low-order bit ‘X1’,so that it outputs the trellis-encoded result as two output bits Z1 andZ0.

FIG. 41( b) illustrates an exemplary case in which a trellis encoder isserially concatenated with the odd component encoder.

In fact, although a plurality of blocks are located between the evencomponent encoder and the trellis encoder, the receiving systemconsiders two blocks to be concatenated with each other, so that itdecodes data. The trellis encoder performs precoding on the high-orderbit ‘X2’ generated from the odd component encoder, and outputs theprecoded result as a most significant bit ‘Z2’. In addition, the trellisencoder performs trellis-encoding on the low-order bit A1′, so that itoutputs the trellis-encoded result as two output bits Z1 and Z0.

FIG. 42 is a trellis diagram including states capable of being acquiredwhen a start state for the even decoder is set to ‘00000’. FIG. 43 is atrellis diagram including states capable of being acquired when a startstate for the odd decoder is set to ‘00000’

For example, if it is assumed that the even component encoder and thetrellis encoder are regarded as a single encoder (i.e., a singleeffective component encoder) in the same manner as in FIG. 41( a), only16 states from among 32 states are effective as shown in FIG. 42. Foranother example, if it is assumed that the odd component encoder and thetrellis encoder are regarded as a single encoder (i.e., a singleeffective trellis encoder) in the same manner as in FIG. 41( b), only 8states are effective as shown in FIG. 43.

In this way, in the case where the component encoder and the trellisencoder are implemented as a single effective component encoder and thenthe encoding of data is carried out in the single effective componentencoder, the number of states to be selected from among 32 states forthe above-mentioned encoding process is changed according to thecomponent encoder structures. In this case, states to be used for theencoding process are changed according to which one of states is used asa start state.

For example, if it is assumed that the odd component encoder and thetrellis encoder are regarded as a single effective component encoder inthe same manner as in FIG. 41( b), the number of states to be used forthe encoding is 8. In addition, if it is assumed that memories of theeffective component encoder shown in FIG. 41( b) are designed to alwaysstart from the state ‘00000’ in the signaling information area, theabove 8 states become ‘00000’, ‘00111’, ‘01010’, ‘01101’, ‘10001’,‘10110’, ‘11011’, and ‘11100’, respectively.

In this way, since only some states from among a total of states areused when the transmission system encodes data of the signalinginformation area, the iterative turbo decoder 7111 of the signalingdecoder 1306 can perform turbo decoding of data using only the effectivestates, thereby greatly reducing complexity of the turbo decoder.

Meanwhile, the derandomizer 7112 performs derandomizing of the iterativeturbo-decoded data, and outputs the derandomized result to thedemultiplexer 7113. The demultiplexer (Demux) 7113 discriminates betweenTPC data composed of 18 bytes and FIC data composed of 51 bytes on thebasis of the derandomized data.

Here, the TPC data is output to the RS decoder 7114 corresponding to anRS (18, 10) of a GF 256. The RS decoder 7114 receives a result of harddecision from the iterative turbo decoder 7111 so as to perform generalRS decoding, or the RS decoder 7114 receives the result of soft decisionfrom the iterative turbo decoder 7111 so as to perform RS erasuredecoding. TPC data (i.e., transmission parameter information)error-corrected by the RS decoder 7114 is output to the operationcontroller 1307. In this case, the RS decoder 7114 further transmits thedecision result to the operation controller 1307, so that it preventsthe occurrence of operational failure which may be generated frommisjudgment of the transmission parameter.

Also, since some information of the TPC data is repeatedly transmittedto each group, decoding performance can be improved using such afeature. For example, in case of FEC mode information such as SCCC orRS, since information of next M/H frame is repeatedly transmitted tothree sub frames at the rear of one M/H frame, even though decoding issuccessfully performed once within the three subframes, there is noproblem in receiving the next M/H frame.

The FIC data discriminated by the demultiplexer 7113 is output to a(TNoG×51) block deinterleaver 7115. The block deinterleaver 7115 is aninverse process of the (TNoG×51) block interleaver of the signalingencoder of the transmitting side.

For example, the (TNoG×51) block interleaver of the transmitting side isa variable-length block interleaver, and interleaves FIC data containedin each subframe in units of a (TNoG (columns)×51 (rows)) block. In thiscase, ‘TNoG’ is indicative of a total number of data groups allocated toa subframe contained in a single M/H frame.

The FIC data block-deinterleaved by the block deinterleaver 7115 isinput to the RS decoder 7116 corresponding to the RS (51, 37) of the GF256. In the same manner as in the RS decoder 7114 for TPC data, the RSdecoder 7116 is able to use both the hard decision value and the softdecision value, and FIC data error-corrected by the RS decoder 7116 isoutput to the FIC processor 1308.

Meanwhile, TNoG value required by the block deinterleaver 7115 can beacquired from the TPC data output from the RS decoder 7114. To this end,the block deinterleaver 7115 includes a controller.

However, since TNoG of next M/H frame is transmitted to three subframesat the rear of one M/H frame, information of TNoG of the currentsubframe may not be obtained through TPC data decoding. For example, ifthe broadcast receiver is turned on at the third subframe (sub-frame #2)and starts to perform FIC decoding to obtain channel information, andperforms FIC block deinterleaving using TNoG within the TPC data, thebroadcast receiver cannot decode the FIC data until it reaches the nextM/H frame.

Accordingly, the present invention suggests a method for decoding FICdata by acquiring TNoG even without using RS-decoded TPC data.

FIG. 44 illustrates a detailed embodiment of a process of extractingTNoG in accordance with the present invention.

The process of acquiring TNoG according to the present invention may beperformed by the signaling decoder 1306, or may be performed by theoperation controller 1307. According to one embodiment of the presentinvention, TNoG is acquired by the signaling decoder 1306. Inparticular, according to one embodiment of the present invention, acontroller is provided at the block deinterleaver 7115 within thesignaling decoder 1306, and acquires TNoG. This is only one example, andthe controller may be provided outside the block deinterleaver 7115.

In other words, if a command to start FIC decoding is input, thesignaling decoder 1306 searches start of next subframe. For example, itis supposed that a command to start FIC decoding is input at the middleof the n−1 th subframe within one M/H frame as shown in (a) of FIG. 44.In this case, start of the nth subframe is searched.

Namely, if the command to start FIC decoding is input, in order toextract start of the subframe, it is identified whether a data groupexists in a corresponding slot. For example, 16 slots are assigned toone subframe. At this time, since known data exist in the data group, itis identified whether the data group exists in the corresponding slot asshown in (b) of FIG. 44 through correlation between a pre-determinedpattern of known data and received data. As another example, informationas to whether the data group exists in the corresponding slot may beprovided from the operation controller 1307.

At this time, if it is identified whether the data group exists in thecorresponding slot, the signaling decoder 1306 performs turbo decoding,signaling derandomizing, and demultiplexing for data of the signalinginformation area within the data group to split TPC data, and performsRS decoding for the split TPC data. Then, the signaling decoder 1306acquires a slot number from the RS-decoded TPC data as shown in (d) ofFIG. 44.

The slot number becomes 0 at a start slot of each sub frame, and has avalue of 15 at the last slot of the corresponding subframe. Accordingly,start of the subframe can be identified by using the slot number.

In other words, the signaling decoder 1306 repeatedly performs thesignaling decoding process until the slot number having a value of 0 isdetected from the TPC data. If the data group within the subframe isassigned and transmitted as shown in (a) of FIG. 44 and the command tostart FIC decoding is input at the middle of the n−1 th subframe, startof the nth subframe is detected through the signaling decoding process.If the start of the subframe is detected, the group counter value isreset to 0.

If start of the nth subframe is detected through the above process, thesignaling decoder 1306 detects a data group from the nth sub frame.

The presence of the data group may be identified using the correlationbetween the known data pattern and the received data, or may be providedfrom the operation controller 1307.

If the data group is detected, the group counter value increases by 1 asshown in (f) of FIG. 44. The turbo decoder 7111 and the derandomizer7112 perform turbo decoding and derandomizing for data of the signalinginformation area within the data group. Subsequently, the demultiplexer7113 performs demultiplexing for the derandomized data to split TPC datafrom FIC data, and the RS decoder 7114 performs RS decoding for thesplit TPC data. The slot number is acquired from the RS-decoded TPCdata. Also, the split FIC data (i.e., 51 bytes) are stored in a buffer(not shown) of the block deinterleaver 7115.

The steps are performed whenever the data group is detected from thesubframe to increase the group counter value by 1, and the buffer of theblock deinterleaver 7115 stores the split FIC data by the demultiplexer7113.

This process is performed until the end of the subframe is detected.According to one embodiment of the present invention, the end of thesubframe is detected using the slot number such as (d) of FIG. 44.According to another embodiment of the present invention, the end of thesubframe is detected using the field synchronizing counter value such as(e) of FIG. 44.

If the end of the subframe is identified, TNoG is calculated using thegroup counter value.

The TNoG value is applied to the FIC data stored in the buffer of theblock deinterleaver 7115 to perform block deinterleaving. The blockdeinterleaved FIC data are input to the RS decoder 7116 and thenRS-decoded by the RS decoder 7116. In case of (g) and (h) of FIG. 44,turbo decoding and derandomizing are performed for the FIC data includedin each data group of the nth sub frame for the nth subframe intervaland then stored in the buffer of the block deinterleaver 7115. The TNoGcalculated is applied to the FIC data of the nth subframe stored in thebuffer of the block deinterleaver 7115 to perform block deinterleaving,RS decoding is performed for the block deinterleaved FIC data.

Meanwhile, the end of the subframe may be detected using either the slotnumber such as (d) of FIG. 44 or the field synchronizing counter valuesuch as (e) of FIG. 44.

In other words, if the slot number acquired from the RS-decoded TPC databecomes 0, it means that a new subframe starts. Accordingly, if the slotnumber becomes 0, it is determined that the previous subframe has ended.In this case, since the group counter value increases by 1, the valueobtained by subtracting 1 from the group counter value becomes the TNoGvalue.

However, next subframe is the first subframe of new M/H frame, and datagroup may not exist in next M/H frame due to PRC. Under thecircumstances, if the end of the subframe is detected using the slotnumber, TNoG cannot be identified until M/H frame where data groupexists is detected, whereby FIC decoding time may be delayed. In thiscase, start of the subframe can be determined using the new slot number,and the number of field synchronization values can be counted toidentify the end of the subframe. This is because that eight fieldsynchronization values in one subframe and field synchronization valuesare transmitted regardless of the presence of the data group. Forexample, if the field synchronization counter value is 8, it isdetermined as the end of the subframe. In this case, the group countervalue becomes the TNoG value. The field synchronization values can alsobe detected through correlation.

FIG. 45 is a diagram showing the form of a known data sequence accordingto an embodiment of the present invention.

FIG. 45( a) shows a long known data sequence. In one embodiment, such along known data sequence may be inserted into a part withoutinterference of main service data in a data group. In a broadcastsystem, known data is inserted into a specific region of a data group,for reception performance improvement. This known data may be used forcarrier synchronization restoration, frame synchronization restorationand channel equalization in a receiver.

FIG. 45( b) shows a segmented known data sequence additionally insertedin an SFCMM.

A broadcast system according to the present invention includes twotransmission modes. In a transmission mode which may be defined as afirst transmission mode or a core mobile mode (CMM), at least 38 packetsof 157 packets which may be transmitted during one slot are reserved formain service data. In a transmission mode which may be defined as asecond transmission mode or a scalable full channel mobile mode (SFCMM),less than 38 packets of 157 packets which may be transmitted during oneslot are reserved for main service data.

That is, the SFCMM refers to a mode in which all or some of VSB datasymbols of a transmission field for broadcast are corrected or extendedfor mobile reception, in the existing system.

In SFCMM, additional known data is inserted into a data group so as toextend a mobile reception region. Additionally provided known data isnot continuously connected due to VSB main data symbol interferenceaccording to a transmission mode and is scattered and segmented. Forexample, known data may be additionally inserted into regions C, Dand/or E of a data group and the known data inserted into these regionsmay be segmented. Additionally inserted known data may have differentpatterns according to transmission modes and the number of unknown VSBdata symbols in a known data period may be changed.

FIG. 46 is a diagram showing the structure of a channel equalizeraccording to an embodiment of the present invention.

A CIR estimator may estimate a channel impulse response (CIR) using aleast squares (LS) method. The LS estimation method obtains a crosscorrelation value p between known data passing through a channel duringa known data period and known data of a receiver and obtains a selfcorrelation value R of the known data. Then, the CIR of a transportchannel is estimated by performing a matrix operation of R⁻¹·p so as toeliminate a self correlation portion present in p which is the crosscorrelation value between the received data and the original known data.

The CIR estimator may estimate the CIR using a least means square (LMS)method. The basic principle of the LMS CIR estimator receives the outputof an unknown transport channel and updates a coefficient value of anadaptive filter such that a difference between the output value of thischannel and the output value of the adaptive filter is minimized

The CIR of a data period other than known data may be estimated usinginterpolation between CIRs obtained in a known data period, and channelequalization is performed by a frequency domain equalizer (FDEQ) so asto obtain equalized data symbols.

As described above, in an SFCMM system, known data which is additionallyprovided is not continuously connected by an unknown VSB main datasymbol according to transmission modes and is scattered and segmented.The VSB main data symbol of the received additional scattered known dataperiod has an unknown value and may not be used as known data.Accordingly, the value of the VSB main data symbol received betweenknown data may be set to 0 and the CIR may then be estimated. However,in this case, a correlation property between the actually receivedunknown VSB data symbol and the set known data may be very bad and thuschannel estimation performance may be deteriorated.

Accordingly, in the embodiment of the present invention, using the basicCIR estimation and FDEQ structure and the iterative structure of adecision device, the value of an unknown VSB data symbol in scatteredknown data additionally provided in the SFCMM is set to a decided symbolso as to substantially secure the length of a training sequence and toeliminate interference of an unknown data symbol, thereby improving CIRestimation performance.

The description of the other configuration of the channel equalizer isreplaced with the description of the structure of the channel equalizerof FIG. 29.

FIG. 47 is a diagram showing the structure of a channel equalizeraccording to another embodiment of the present invention.

The channel equalizer according to another embodiment of the presentinvention includes a channel equalizer 47100 having a CIR estimation andFDEQ structure and/or a decision device 47200.

The description of the channel equalizer 47100 is replaced with theabove description of the channel equalizers.

The decision device 47200 may decide data (VSB data) included betweensegmented known data sequences using equalizer output and an inputerror-corrected by an FEC aided-device. A newly estimated known datasequence decided by the decision device 47200 is used as an input of aknown data sequence necessary for CIR estimation of a next iteration.

A channel equalization method according to another embodiment of thepresent invention will now be described.

Symbols passing through the channel equalizer 47100 including a CIRestimator and an FDEQ are input to the decision device 47200. The VSBdata symbol located in the additionally provided scattered known dataperiod is used to estimate and decide the transmitted symbol in thedecision device 47200. The decision device 47200 may use hard decision.The decision value used for decision may be set in four or eight steps.As necessary, the decision value may be weighted. The signal input tothe decision device 47200 may use the output of the channel equalizer asdescribed above and the accuracy of symbol decision using the symbolerror-corrected by FEC may be improved to obtain improved performance. Aknown data sequence which will be used in a next iteration uses a knownpattern without change in a scattered known data period and is replacedwith an estimated symbol only at the position of the unknown VSB symbol,thereby configuring a known data sequence.

In case of an equalizer output symbol passing through an initial channelequalizer, channel estimation performance is deteriorated due tointerference of an unknown data symbol in known data and an SNR of theequalizer output is bad. However, a plurality of channel equalizationorders is iterative such that a large number of unknown VSB data isdecided as an actual transmission symbol. Thus, CIR estimation andchannel equalization performance can be improved.

FIG. 48 is a diagram showing an iterative channel equalizer according toan embodiment of the present invention.

As shown in FIG. 48, iterative channel equalizers 48120 and 48220 anddecision devices 48130 and 48230 are connected and such a connectionrelationship is conceptually repeated. FIG. 48( a) is a conceptualdiagram of the serialized structure of steps. Delay buffers 48110 and48210 are necessary for CIR estimation, FDEQ and decision of a newtraining sequence. A symbol input to the channel equalizers is subjectedto N iterations so as to obtain a channel-equalized symbol.

FIG. 48( b) shows a structure in which a CIR estimator 48320 and adecision device 48330 are configured and a data symbol input to achannel equalizer necessary for each iteration and a new known datasequence decided in a previous iteration are selected as an input. Theoutput of the channel estimator 48320 is connected to the output of thedecision device 48330 up to an (N−1)-th iteration and an equalizedsymbol is output in an N-th iteration.

In the iterative channel equalizer, the input of the decision device maybe configured by receiving the output of the symbol error-corrected byFEC in addition to the output of the channel equalizer. At this time, incase of the delay buffer, a delay time necessary to reconfigure theoutput of the error-corrected symbol must be considered.

FIG. 49 is a diagram showing a channel equalizer including apre-equalizer and a post equalizer according to an embodiment of thepresent invention.

The channel equalizer including the pre-equalizer and the post equalizeraccording to the embodiment of the present invention may include a delaybuffer 49010, a pre-equalizer 49020, a decision device 49030 and/or apost equalizer 49040 (CIR estimator/FDEQ).

The delay buffer 49010 serves to buffer an input signal in considerationof a delay time required for processing of a signal in the pre-equalizer49020 and the decision device 49030. At this time, if an error-correctedsymbol is used as an input of the decision device 49030, the size of thebuffer may be decided in consideration of the delay time.

The pre-equalizer 49020 performs a pre equalization process in order toprevent performance from being deteriorated in a known data period dueto unknown VSB data, before channel equalization is performed by thepost equalizer 49040. The pre-equalizer 49020 serves to configure a newknown sequence estimated by reducing decision error with respect to anunknown VSB data symbol located in the scattered known data periodthrough an iterative convergence process in the additionally providedscattered known data period.

The decision device 49030 serves to estimate the unknown VSB data symbollocated between scattered known data. For example, the decision device49030 receives the output of the pre-equalizer, performs errorcorrection by FEC and decides a CSB data symbol based on theerror-corrected symbol.

The decision value used for decision may be set in four or eight steps.As necessary, the decision values may be weighted. A known data sequenceto be used in the post equalizer 49040 uses a known pattern withoutchange in a scattered known data period and is replaced with anestimated symbol only at the position of the unknown VSB symbol, therebyconfiguring a known data sequence.

The post equalizer 49040 estimates a CIR of an input signal,approximates the CIR, and performs channel equalization using an FDEQ.The post equalizer 49040 approximates the CIR using interpolation in adata period other than the estimated known data sequence period. Thepost equalizer 49040 may receive a new known data sequence processed bythe pre-equalizer 49020 and the decision device 49030 and performchannel equalization. The post equalizer 49040 may have the structure ofthe above-described channel equalizer.

As described above, in the present invention, a pre-equalization processis performed with respect to additional scattered known data by thepre-equalizer 49020 at a previous stage of the post equalizer 49040, anda new training sequence processed by the decision device 49030 is inputto the post equalizer, thereby improving channel equalizationperformance.

FIG. 50 shows the structure of a decision feedback equalizer of apre-equalizer according to an embodiment of the present invention.

The pre-equalizer may include a channel equalizer which is a decisionfeedback equalizer of FIG. 46. The decision feedback equalizer includesa feed-forward filter and a feed-back filter and updates a filtercoefficient using decision errors. A convergence time necessary forfilter coefficient convergence of the decision feedback equalizer may beset to a predetermined value and the update process may be finished ifan output SNR of the pre-equalizer is equal to or greater than apredetermined threshold.

FIG. 51 is a diagram showing the structure of a channel equalizeraccording to another embodiment of the present invention.

As shown in FIG. 51, the channel equalizer may include a frequencyregion converter 51010, a distortion compensator 51020, a time regionconverter 51030, a first coefficient calculator 51040, a secondcoefficient calculator 51050 and/or a coefficient selector 51060.

The frequency region converter 51010 may include an overlap unit 51011and/or a first FFT unit 51012.

The time region converter 51030 may include an IFFT unit 51031 and/or asave unit 51032.

The first coefficient calculator 51040 may include a CIR estimator51041, an interpolator 51041, a second FFT unit 51043 and/or acoefficient calculator 51044.

The second coefficient calculator 51050 may include a decider 51051, aselector 51052, a subtractor 51053, a zero padding unit 51054, a thirdFFT unit 51055, a coefficient update unit 51056 and/or a delay unit51057.

At this time, the coefficient selector 51060 may include a multiplexer(MUX) for selecting input data depending on whether current input datais data of the region A/B or the region C/D or E. That is, thecoefficient selector 51060 selects the equalization coefficient of thefirst coefficient calculator 51040 if the input data is data of theregion A/B and selects the equalization coefficient of the secondcoefficient calculator 51050 if the input data is data of the region C/Dor E.

In the channel equalizer according to the embodiment of the presentinvention, the received data is input to the overlap unit 51011 of thefrequency region converter 51010 and the first coefficient calculator51040. The overlap unit 51011 overlaps the input data according to apredetermined overlap ratio and outputs the overlap data to the firstFFT unit 51012. The first FFT 51012 transforms the overlap data of atime region into overlap data of a frequency region through FFT andoutputs the overlap data of the frequency region to the distortioncompensation unit 51020 and the delay unit 51057 of the secondcoefficient calculator 51050.

The distortion compensation unit 51020 complex-multiplies the overlapdata of the frequency region output from the first FFT unit 51012 by anequalization coefficient output from the coefficient selector 51060 tocompensate for channel distortion of the overlap data output from thefirst FFT 51012 and outputs the compensated data to the IFFT unit 51031of the time region converter 51030. The IFFT 51031 IFFTs the overlapdata, channel distortion of which is compensated for, to transform theoverlap data into the overlap data of the time region and outputs theoverlap data of the time region to the save unit 51032. The save unit51032 extracts only valid data from the overlap data of the time regionsubjected to channel equalization, outputs the extracted valid data fordata decoding and outputs the valid data to the second coefficientcalculator 51050 for coefficient update.

The CIR estimator 51041 of the first coefficient calculator 51040estimates a CIR using data received during a known data period andreference known data generated by a receiver and outputs the CIR to theinterpolation unit 51042. The CIR estimator 51041 estimates the CIRusing data received during a field synchronization period and referencefield synchronization data generated by the receiver and outputs the CIRto the interpolation unit 51042.

The interpolation unit 51042 estimates CIRs located between theestimated CIRs, that is, CIRs of a period in which known data is notpresent, using the input CIRs by a predetermined interpolation methodand outputs the estimated result to the second FFT 51043. The second FFTunit 51043 transforms the input CIRs into the CIRs of the frequencyregion and outputs the CIRs to the coefficient calculator 51044. Thecoefficient calculator 51044 calculates a frequency region equalizationcoefficient satisfying a condition for minimizing mean square errorusing the CIRs of the frequency region and outputs the frequency regionequalization coefficient to the coefficient selector 51060.

The decider 51051 of the second coefficient calculator 51050 selects adecision value closest to the equalized data from among a plurality ofdecision values, for example, eight decision values and outputs theselected decision value to the selector 51052. The selector 51052selects the decision value of the decider 51051 in a general dataperiod, selects known data in a known data period, and outputs theselected decision value or known data to the subtractor 51053. Thesubtractor 51053 subtracts the output of the time region converter 51030from the output of the selector 51052 to obtain an error value andoutputs the error value to the zero padding unit 51054.

The zero padding unit 51054 pads zero to the input error value by theoverlap amount of the received data and outputs the error value, towhich zero is padded, to the third FFT unit 51055. The third FFT unit51055 transforms the error value of the time region, to which zero ispadded, into the error value of the frequency region and outputs theerror value of the frequency region to the coefficient update unit51056. The coefficient update unit 51056 updates a previous equalizationcoefficient using the delayed data of the frequency region and the errorvalue of the frequency region and outputs the updated equalizationcoefficient to the coefficient selector 51060. At this time, the updatedequalization coefficient is stored to be used as a previous equalizationcoefficient in the future.

The coefficient selector 51060 selects the equalization coefficientcalculated by the first coefficient calculator 51040 if the input datais data of the region A/B, selects the equalization coefficient updatedby the second coefficient calculator 51050 if the input data is data ofthe region C/D or E, and outputs the selected equalization coefficientto the distortion compensation unit 51020.

FIG. 52 is a diagram showing a CIR estimator according to an embodimentof the present invention.

Referring to FIG. 52, the CIR estimator includes delay units T forsequentially delaying output data {circumflex over (x)}^((n)) of aprevious stage of the CIR estimator, multipliers for multiplying theoutput data of the delay units T by error data e(n) and coefficientupdate units for updating coefficients by the outputs of themultipliers, the number of which is equal to the number of tabs.

For convenience of description, the multipliers, the number of which isequal to the number of tabs, are referred to as a first multiplier unit.Multipliers for multiplying the output data of the previous stage of theCIR estimator and the output data of the delay units T (excluding theoutput data of the last delay unit) by the output data of thecoefficient update units are further included, the number of which isequal to the number of tabs. For convenience of description, thesemultipliers are referred to as a second multiplier unit. An adder foradding all the output data of the multipliers of the second multiplierunit and outputting an equalizer input estimation value ŷ^((n)) and asubtractor for outputting a difference between the output ŷ^((n)) of theadder and equalizer input data y(n) as error data e(n) are furtherincluded.

The decision value of the equalized data is input to the first delayunit and the first multiplier of the second multiplier unit in the CIRestimator in a general data period and the know data is input to thefirst delay unit and the first multiplier of the second multiplier unitin the CIR estimator in the known data period.

The input data {circumflex over (x)}^((n)) is sequentially delayedthrough the delay units i corresponding in number to the number of tabsand connected in series. The output data of the delay units and theerror data e(n) are multiplied by the multipliers of the firstmultiplier unit to update the coefficients of the coefficient updateunits. The updated coefficients of the coefficient update unit aremultiplied by the input data {circumflex over (x)}^((n)) and the outputdata of the delay unit excluding the last delay unit by the multipliersof the second multiplier unit and the multiplied coefficients are inputto the adder. The adder adds all the output data of the multipliers ofthe second multiplier unit and outputs the equalizer input estimationvalue ŷ^((n)) to the subtractor. The subtractor outputs the differencebetween the estimation value ŷ^(n) and the equalizer input data y(n) tothe multipliers of the first multiplier unit as the error data e(n). Atthis time, the error data e(n) is output to the multipliers of the firstmultiplier unit through the delay units T.

The coefficient of the filter is continuously updated using theabove-described process and the outputs of the coefficient update unitsbecome the CIR outputs of the CIR estimator in every FFT period.

The CIR estimator according to the present invention may estimatechannel impulse response (CIR) using a least squares (LS) method. The LSestimation method obtains a cross correlation value p between known datapassing through a channel during a known data period and known data of areceiver and obtains a self correlation value R of the known data. Then,the CIR of a transport channel is estimated by performing a matrixoperation of R⁻¹·p so as to eliminate a self correlation portion presentin p which is the cross correlation value between the received data andthe original known data.

In addition, the CIR estimator may estimate a CIR using a least meanssquare (LMS) method. The basic principle of the LMS CIR estimatorreceives the output of an unknown transport channel and updates acoefficient value of an adaptive filter such that a difference betweenthe output value of this channel and the output value of the adaptivefilter is minimized. When transmission data is x(n), a transport channelmay be modeled to a finite impulse response (FIR) filter in which animpulse response is h(n) and an adder for adding a noise component n(n).When the output of this channel is input to an equalizer, an equalizerinput y(n) is expressed as follows.

${y(n)} = {{{{\overset{->}{x}(n)}*{\overset{->}{h}(n)}} + {n(n)}} = {{\sum\limits_{k = 0}^{k = {L - 1}}\; {{x\left( {n - k} \right)}{h(k)}}} + {n(n)}}}$

In this equation, “*” denotes a convolution operation and L denoteschannel length.

The LMS channel estimator configures an equalizer input estimation valueŷ^((n)) using a discrimination value {circumflex over (x)}^((n)) and acoefficient {right arrow over (w)}^((n)) of the equalizer as follows.

${\hat{y}(n)} = {{{\overset{\overset{->}{\hat{}}}{x}(n)}*{\overset{->}{w}(n)}} = {\sum\limits_{k = 0}^{k = {L - 1}}\; {{\hat{x}\left( {n - k} \right)}{w(k)}}}}$

The filter coefficient {right arrow over (w)}^((n)) is updated using thedifference between the equalizer input and the equalizer inputestimation value as errors.

e(n)=y(n)−{circumflex over (y)}(n)

w(i+1)=w(i)+μ·e(n)·x(n), (i=0, 1, . . . , L−1)

In this equation, μ denotes a step size of errors.

If known data is used for LMS CIR estimation, since the receiver isalready aware of the value of the data, instead of the discriminationvalue {circumflex over (x)}^((n)) of the equalizer, the transmissiondata value x(n) may be used. An iterative CIR may be estimated withrespect to one known data region.

When the CIR estimator obtains a cross correlation value between thereceived data and the original known data using a correlator, anapproximate CIR is obtained and this information may be used for LMS CIRestimation. It is assumed that a channel tab having a highestcorrelation value among cross correlation values is referred to as amain tab and a middle value between the position of an earliest channeltab (pre tab) and the position of a latest channel tab (post tab) isreferred to as a channel center point. At this time, it is assumed thata value obtained by subtracting the position of the main tab from theposition of the channel center point is referred to as a channel centermovement value s. This value may have a positive value or a negativevalue.

The methods of obtaining the CIR for channel equalization in each regionof the data group, which have been described up to now, are onlyexemplary and may be variously applied, although the present inventionis not limited to the above-described embodiments.

FIG. 53 shows a CIR estimation method of a known data region accordingto an embodiment of the present invention.

The LMS CIR estimation method of the known data region shown in FIG. 53will now be described. In iterative LMS CIR estimation, known data or avalue of 0 is input to x(n) and received data is input to y(n). When thelength of known data is N, received data corresponding to a first partof the known data region is y_(k)(0) and an input corresponding to alast part thereof is y_(k)(N−1). With respect to any positive integer a,y_(k)(−a) denotes a value which is input earlier than y_(k)(0) by a andy_(k)(N−1+a) denotes a value which is input later than y_(k)(N−1) by a.First data of known data is x_(k)(0), last data is x_(k)(N−1), andx_(k)(−a) and x_(k)(N−1+a) have a value of 0.

ŷ _(k) =w(0)·x _(k)(L/2−s+n)+w(1)·x _(k)(L/2−s−1+n)+ . . . +w(L−1)·x_(k)(L/2−s−L+1+n)

ŷ_(k)(n) is defined by the above equation and a filter coefficient isupdated using a difference between ŷ_(k)(n) and y_(k)(n) as errors. Thisoperation is performed from n=0 to n=N−1 and is performed again at n=0.CIR estimation performance may be changed according to a method ofsetting the range of n.

FIG. 54 is a diagram showing a CIR estimation method of a known dataregion according to another embodiment of the present invention.

FIG. 54 is a conceptual diagram when performing LMS CIR estimation usingthe CIR estimation method of FIG. 53 from n=s to n=N−1+s. The CIRestimation method is described above with reference to FIG. 53 and adetailed description thereof will thus be omitted.

FIG. 55 is a diagram showing a CIR estimation method of a known dataregion according to another embodiment of the present invention.

The CIR estimation method is described with reference to FIGS. 53 and54. CIR estimation of FIG. 55 is narrower than that of FIG. 54.

For example, a combination of the method of FIG. 54 and the method ofFIG. 55 may be used. For example, in iterative LMS CIR, CIR estimationis first performed several times using the method of FIG. 54 and is thenperformed several times using the method of FIG. 55. An approximate CIRvalue obtained using a correlator may be used as an initial value of afilter coefficient.

FIG. 56 is a diagram showing a method of selecting a sparse window forCIR measurement according to an embodiment of the present invention.

In CIR estimation using an LMS method, a sparse LMS CIR estimationmethod of selectively discriminating CIR equal to or greater than acertain threshold and updating a filter coefficient in which the lengthof the selected channel is less than L is applicable. The sparse LMS CIRestimation method updates only the filter coefficient corresponding to avalid CIR in which the length is less than the total length L, therebyimproving a convergence speed. In addition, noise interference in aninvalid period is eliminated to increase accuracy of CIR estimation.

In order to perform the sparse LMS CIR estimation method, initial CIRinformation necessary to obtain the position of a valid sparse CIRwindow is necessary. The necessary CIR may be obtained by the followingembodiment.

First, an output value of cross correlation between original known data,of which a receiver is aware, and received data is divided by an averagepower intensity of y_(k)(n) and normalized to obtain approximate CIRinformation. The CIR information may be computed using the followingequation.

${{{CIR}_{initial}(n)} = {\frac{{y_{k}(n)}*{x(n)}}{E\left\lbrack {{y_{k}(n)}}^{2} \right\rbrack} = \frac{\sum\limits_{l = 0}^{L_{PN} - 1}\; {{y_{k}\left( {n - l} \right)}{x(l)}}}{E\left\lbrack {{y_{k}(n)}}^{2} \right\rbrack}}},\left( {{n = 0},1,\ldots \mspace{14mu},{L - 1}} \right)$L_(PN)

where, L_(pn) denotes the length of the known data.

Second, in a current known data period, a filter coefficient w(n)^(m) bywhich LMS CIR estimation is performed with respect to a previous tabhaving a channel length of L is used as approximate CIR information upto m initial iterations.

CIR_(Initial)(n)=w(n)^(m), (n=0, 1, . . . , L−1)

Third, final CIR information estimated in a previous known data periodmay be stored and used.

If a position where the power level of a CIR exceeds a certain thresholdλ_(Threshold) is p, a period of p±q centered on P is discriminated as avalid CIR region. Finally, a sum of detected windows of a plurality ofCIRs exceeding the threshold is decided as a sparse window.

In FIG. 56( a), the total length of the channel is L, valid channelinformation is located at a part of the total period, and errors may becaused due to noise present in the whole period in a process ofobtaining ŷ_(k)(n).

In FIG. 56( b), a valid CIR is selected in a region of ±q centered on pwith respect to the position of the CIR exceeding a certain threshold.

Finally, as shown in FIG. 56( c), a sparse window may be decided by asum of detected windows.

The decided sparse window is mapped to the filter coefficientcorresponding to CIR_(initial)(n) and a filter coefficient correspondingto an invalid region may be set to “0”. An initial value of a sparse LMSCIR estimation method may be set to a filter coefficient to which awindow is mapped, and a filter coefficient to which a window is mappedis updated and converged using an iterative method.

As to the LMS CIR estimation method, a convergence speed and a jitterproperty after convergence are applied according to a step size. At thistime, the range of a converged step size is related to the number ofselected filter coefficients. If the number of valid filter coefficientsis reduced using the sparse LMS CIR estimation method, convergence israpidly performed by increasing the allowed step size. For example, inan iterative convergence process, the step size is initially increasedto achieve fast convergence and is then decreased to stabilize thejitter property after convergence.

FIG. 57 is a diagram showing a sparse LMS CIR estimator according to anembodiment of the present invention.

The CIR estimator of FIG. 57 has a structure similar to that of theestimator of FIG. 52. The sparse LMS CIR estimator of FIG. 57 receives aCIR having a channel length of L, which is obtained by theabove-described method, as an input of a sparse CIR window controllerand decides the position of a valid CIR using the received CIR.

A CIR obtained in a known data region included in the structure of thedata group shown in FIG. 4 interpolated with a region other than knowndata in the region B4, B5, B6 and/or B7 so as to estimate the CIR ofthis region.

At this time, if the sparse LMS CIR estimation method is used, errorrate may be increased in the interpolation process due to a differencebetween sparse windows estimated in the previous and next known dataregions.

FIG. 58 is a diagram showing generation of interpolation error due to asparse window according to an embodiment of the present invention.

Referring to FIG. 58, in case of a CIR in known data regions of (t−1)and t, a middle tab is equal to or less than a certain threshold, isregarded as an invalid CIR and is excluded from the sparse window so asnot to influence performance in the interpolation period. However, as toa CIR in a known data region of (t+1), a middle tab is greater than thethreshold and is regarded as a valid CIR, and interpolation error may becaused in the interpolation period due to discontinuity of the selectedsparse windows in the known data regions of t and (t+1). Accordingly, inorder to minimize error of the interpolation period, the selected sparsewindows in the adjacent known data periods must have continuity. FIG. 9shows an embodiment of using information about adjacent sparse windowsfor minimizing interpolation error.

FIG. 59 is a diagram showing a method of using information aboutadjacent sparse windows for minimizing error in an interpolation periodaccording to an embodiment of the present invention.

In the initial CIR information obtained by the above-described methods,the sparse window information obtained in the known data regions of t−1,t and t+1 shown in FIG. 58 are respectively referred to as WIN_(t−1),WIN_(t) and WIN_(t+1). In order to minimize interpolation error, thefinal sparse window WIN_(t) ^(f) of the known data region of t may beacquired using an OR operation such that discontinuity between thesparse window WIN_(t−1) and WIN_(t+1) of t−1 and t+1 corresponding tothe previous and next known data regions is removed.

WIN_(t) ^(f)=(WIN_(t−1)) OR (WIN_(t)) OR (WIN_(t+1))

As to first or last known data, sparse window information of one pieceof adjacent known data may be used.

A filter coefficient corresponding to the final sparse window WIN_(t)^(f) is updated as an initial value using the CIR obtained by theabove-described methods and an iterative LMS CIR estimation method isperformed.

In the iterative LMS CIR estimation process described in FIGS. 53 and54, the filter coefficient is updated in order from y_(k)(0) toy_(k)(N−1) or from y_(k)(s) to y_(k)(N−1+s) according to the input orderof known data and the process of updating the filter coefficient at theposition corresponding to first input data is repeated in the nextiteration. However, repeating the process of updating the filtercoefficient at the position of the first data in next iteration requiresa process of filling a known data sequence buffer {right arrow over(x)}_(k)(n) for obtaining ŷ_(k)(n) with new data and thus excess timedelay occurs. In order to reduce excess time delay, in first iteration,the filter is sequentially updated from y_(k)(0) to y_(k)(N−1) or fromy_(k)(s) to y_(k)(N−1+s) and, in the next iteration, the filter isupdated in reverse order, that is, from y_(k)(N+1) to y_(k)(0) or fromy_(k)(N−1+s) to y_(k)(s), such that the input value in the known datasequence buffer {right arrow over (x)}_(k)(n) is changed one by one.Thus, it is possible to reduce time necessary to correct the buffer periteration.

FIG. 60 is a flowchart illustrating a method of receiving and processinga broadcast signal at a receiver according to an embodiment of thepresent invention.

A tuner of a broadcast receiver receives a DTV signal including a datagroup (s60010). The received data group includes mobile service data,segmented known data sequences, long known data sequences and/ortransmission parameter data. In the segmented known data sequences andthe long known data sequences, as shown in FIGS. 8 to 17, long knowndata connected to exceed one segment of the data group may be defined asa long known data sequence and short known data scattered in one segmentmay be defined as a segmented known data sequence. This process may beperformed by the channel synchronizer or the antenna unit of thereceiver shown in FIG. 22.

The broadcast receiver compensates for a carrier frequency offset of theDTV signal (s60020). The broadcast receiver may compensate for thecarrier frequency offset by multiplying the carrier recovered by thecarrier recovery block of the channel synchronizer by an input signal.For example, this process may be performed by the channel synchronizerof the receiver shown in FIG. 22.

The broadcast receiver performs channel equalization with respect to theDTV signal, in which the carrier frequency offset is compensated for,using at least one known data sequence of the segmented known datasequences and the long known data sequences included in the data group(s60030). The channel equalization process may include the techniquesdescribed above in association with the channel equalizer. For example,the channel equalization process may be used to perform forward errorcorrection (FEC) decoding with respect to data located between thesegmented known data sequences and estimate a channel impulse response(CIR) using the FEC-decoded data as known data. This process may beperformed by the channel equalizer of the receiver shown in FIG. 22.

As described above, the transmitting system, the receiving system, andthe method of processing broadcast signals according to the presentinvention have the following advantages.

When transmitting mobile service data through a channel, the presentinvention may be robust against errors and backward compatible with theconventional digital broadcast receiving system.

This invention extends a region for mobile service data in a slot. Thus,the transmitter can transmit more mobile service data.

This invention has an advantage enhancing the reception performance of abroadcast signal at a reception system, and a method for processing abroadcast signal by inserting additional known data in regions C, D andE.

When performing block deinterleaving on the turbo-decoded FIC data, byacquiring TNoG using slot information and data group information, thepresent invention may decode the FIC data, even when the TNoGinformation of the current subframe cannot be acquired by TPC datadecoding. Therefore, the decoding performance of the FIC data may beenhanced.

After performing CRC-RS decoding, a CRC syndrome check may be performedonce again on the RS frame, and the checked result is marked on anerror_indicator field of the M/H service data packets configuring thepayload of the RS frame. Thus, the likelihood of malfunction occurringin the block receiving and processing the M/H service data packet may bereduced, thereby enhancing the overall performance of the receivingsystem.

According to the embodiments of the present invention, it is possible toperform channel equalization using segmented known data sequences.

According to the embodiments of the present invention it is possible toshorten a time required for performing channel equalization.

Finally, the present invention is even more effective when applied tomobile and portable receivers, which are also liable to a frequentchange in channel and which require protection (or resistance) againstintense noise.

It will be apparent to those skilled in the art that variousmodifications and variations can be made in the present inventionwithout departing from the spirit or scope of the inventions. Thus, itis intended that the present invention covers the modifications andvariations of this invention provided they come within the scope of theappended claims and their equivalents.

1-10. (canceled)
 11. A method for transmitting broadcast signals, themethod comprising: first encoding mobile data for a mobile service;second encoding the first encoded mobile data; forming a data unitincluding the second encoded mobile data; and transmitting the broadcastsignals including a signal frame, wherein the signal frame includesmobile service data symbols carrying the second encoded mobile data inthe formed data unit, wherein a number of mobile service data symbols isscalable, wherein the data unit includes scattered known data andtransmission parameter data.
 12. The method of claim 11, wherein thetransmission parameter data includes version information representing aversion of a structure of the transmission parameter data.
 13. Themethod of claim 11, wherein the transmission parameter data includesinformation related to the size of the formed data unit.
 14. The methodof claim 11, the formed data unit further includes FIC (Fast InformationChannel) carrying cross-layer information.
 15. An apparatus method fortransmitting broadcast signals, the apparatus comprising: a firstencoder for first encoding mobile data for a mobile service; a secondencoder for second encoding the first encoded mobile data; a formatterfor forming a data unit including the second encoded mobile data; and atransmitter for transmitting the broadcast signals including a signalframe, wherein the signal frame includes mobile service data symbolscarrying the second encoded mobile data in the formed data unit, whereina number of mobile service data symbols is scalable, wherein the dataunit includes scattered known data and transmission parameter data. 16.The apparatus of claim 15, wherein the transmission parameter dataincludes version information representing a version of a structure ofthe transmission parameter data.
 17. The apparatus of claim 15, whereinthe transmission parameter data includes information related to the sizeof the formed data unit.
 18. The apparatus of claim 15, the formed dataunit further includes FIC (Fast Information Channel) carryingcross-layer information.